System for Operating Dc Motors and Power Converters

ABSTRACT

A system is disclosed for driving a DC motor ( 15 ) under conditions of a controlled average current. An inductive element may be arranged for connection in series with the DC motor. A switch ( 14 ) is preferably coupled to the inductive element for connecting and disconnecting a terminal of the inductive element from the voltage source. A diode may be arranged for connection in parallel with a combination of the inductive element and the DC motor arranged in series, with the appropriate polarity so that current circulating through the inductive element circulates through the diode when the switch disconnects the terminal from the voltage source. A capacitor is arranged for connection in parallel with the motor, for limiting a resulting voltage over the motor or for storing charge depending on the embodiment of the invention. A device for measuring a current through the motor is provided, and a device ( 13 ) for controlling operation of the switch dependent upon the measured current in the motor is also provided. An airflow apparatus is also disclosed.

FIELD OF THE INVENTION

The present invention relates generally to motors and power suppliesand, in particular, systems for opening a direct-current (DC) motor orpower converters.

BACKGROUND

In the treatment of steep apnoea and other respiratory disorders, apositive air pressure is used applied to the patient airway. Theequipment used is known as an air flow-generator.

A method used to generate air pressure is shown in FIG. 1. A brush-lessDC motor (16) is used to drive a turbine or blower (15). The turbine(15) generates the airflow w for the patient. The brush-less DC motorcontroller (14) in conjunction with the control electronics of the flowgenerator (13), receive power from a power supply (12) that is connectedto the AC main through a filter (11). Sometimes, the filter is builtinto the power supply itself. Control signals are sent from the controlelectronics (13) to the brush-less DC motor controller (14), so thespeed of the motor (16) can be controlled.

The pressure and the amount of air delivered depend on the speed of theturbine. In some types of equipment, pressure and flow sensors are usedto monitor these variables and change the speed of the motor to achievethe desired effect. Also, in some cases, the speed of the motor ischanged, alternating between a high and a low value, either in responseto the patient respiration or as part of an automatic cycle. Equipmentperforming in this way is known as bi-level devices.

In FIG. 1, a single power supply (12) provides power to both, the motordriving circuits (14) and the control electronics (13).

A brush-less DC motor, or BLDCM, is a DC motor, with an electroniccommutator. FIG. 2 shows a block diagram of one type of permanent magnetbrush-less DC motor along with its electronic commutator.

The driving electronics consist of a logic circuit (22), that control aset of electronic switches (21) that switch power to the motor windings(23) much as the brushes do in a conventional DC motor. Current throughthe windings (23) generates forces in the rotating magnets (24), makingthe rotor of the motor to spin. The switches (21) can connect the end ofits corresponding winding to either the positive or the negative side ofthe DC voltage source, and also they can leave the winding unconnected.

The logic circuit (22) of the electronic commutator has as an output twocontrol signals per switch, shown in the figure as signal groups SWC1 toSWC3, of two lines each. The motor has hall-effect sensors (25, 26 and27) that are used by the logic circuit (22) to detect the position ofthe rotor and switch the right waveforms to the windings (23).Typically, the industry uses a three phase motor (three windings) thatare depicted in Y-configuration, for example, but may also be in atriangle configuration.

As the axis of the motor rotates, the motor windings are driven withthree trapezoidal 6-step waveforms. During each step, voltage is appliedto two windings only.

There is also a sensor-less mode of operation, in which a specialcontroller monitors the voltage in the winding that has been left opencircuit to read the back-emf generated in the winding as the motor axisrotates.

In a CPAP application, like the one shown in FIG. 1, the BLDCM (16)takes considerable power especially during its acceleration periods. Ina typical CPAP application, a motor can take around two amps at 24 volts(or more if a 12 volts motor is used), depending on the pressure andflow generated, and the particular motor chosen. The electronicsnecessary to perform the control, however, can be designed so theelectronics take under 0.1 amps of current at a relatively low voltage.Most of the electronics can work with 5 volts while only the pressureand flow sensors may need more, depending on the implementation.

The power supply for the motor (16 in FIG. 1) and the switching sectionof the electronic commutator (21 in FIG. 2) require more relaxedspecifications than the power supply for the control electronics (13 inFIG. 1) or the electronic commutator logic (22 in FIG. 2). A motor is aforgiving load for a power supply. As the motor's to mechanicalcharacteristics work as a low pass filter, the motor can tolerate arelatively large ripple voltage. In fact, the ripple can be up to 100%without affecting operation. Furthermore, some applications of motors(e.g., driving a fan or turbine) can tolerate the discontinuous torquethat comes with a discontinuous supply of current.

A brushless-dc-motor-driven ventilation fan shares most of the buildingblocks of an air flow generator for medical applications. The maindifferences are:

-   -   The mechanical design of the turbine or fan itself, since a flow        generator needs to produce more pressure.    -   The ventilation fan, normally, does not interface with flow and        pressure sensors. Thus, the control electronics of a ventilation        fan are simpler and should draw less current.

Regulations like the European Standard EN 60555 and the InternationalStandard IEC 555-2 limit the current harmonic content of mains suppliedequipment. This requirement applies to both the medical application ofthe DC motors and the ventilation fans. Power factor correction must betaken into account for all new designs. Power factor correction can add20 to 30% to the cost of the power supply of equipment (see Reference 10in Appendix C). Hence, there can be a relatively substantial saving inthe cost of the equipment if the function of power factor correction isintegrated with the DC motor driver for equipment working from the ACmains.

SUMMARY

In accordance with an aspect of the invention, there is provided asystem for driving a direct-current (DC) motor under conditions of acontrolled average current, the system comprising an inductive elementfor connection in series with the DC motor; a first switch coupled tothe inductive element for connecting and disconnecting a terminal of theinductive element remote from the DC motor to a voltage source; a secondswitch connected in parallel with a combination of the inductive elementand the DC motor arranged in series, controlled so that a currentcirculating through the inductive element circulates through the secondswitch if the first switch disconnects the terminal of the inductiveelement from the voltage source; a capacitor arranged for connection inparallel with the DC motor to limit a resulting voltage over the DCmotor, means for measuring a current through the DC motor; and means forcontrolling operation of the first and second switches dependent uponthe measured current in the DC motor.

In accordance with another aspect of the invention, there is provided asystem for driving a direct-current (DC) motor under conditions ofcontrolled average current, the system comprising a capacitor arrangedfor connection in parallel with the motor to limit a resulting voltageover the motor, the other terminal of the parallel combination of thecapacitor and the motor connected to a common terminal; an inductiveelement connected to the common terminal; a first switch coupled to theinductive element for connecting and disconnecting a terminal of theinductive element to a voltage source; a second switch connected inseries with the parallel combination of the motor and the capacitor, andconnected to the common node between the first switch and the inductiveelement, controlled so that the current circulating through theinductive element circulates through the second switch if the firstswitch disconnects the terminal from the voltage source; means formeasuring a current through the motor, and means for controllingoperation of the first and second switches dependent upon the measuredcurrent in the motor.

In accordance with yet another aspect of the invention, there isprovided a system for driving a direct-current (DC) motor underconditions of a controlled average current, the system comprising acapacitor arranged for connection in parallel with the motor to limit aresulting voltage over the motor, a terminal of the capacitor and themotor being connected to a DC voltage source; an inductive elementconnected to a common node of the DC voltage source, the capacitor andthe motor; a first switch coupled to the inductive element forconnecting and disconnecting a terminal of the inductive element to theterminal of the voltage source not connected to the parallel combinationof the capacitor and the motor; a second switch connected in series withthe parallel combination of the motor and the capacitor and connected tothe common node between the first switch and the inductive element,controlled so that the current circulating through the inductive elementcirculates through the second switch if the first switch disconnects theterminal from the voltage source; means for measuring a current throughthe motor; and means for controlling operation of the first and secondswitches dependent upon the measured current in the motor.

In accordance with a further aspect of the invention, there is provideda system for driving a direct-current (DC) motor under conditions of acontrolled average current, a voltage of a DC power supply having alarger or smaller value than a motor nominal voltage, the systemcomprising an inductive element for connection in series with the DCmotor; an arrangement including a plurality of switches, diodes and amagnetic system, the arrangement coupled to the inductive element forconnecting and disconnecting a terminal of the inductive element remotefrom the motor to a voltage source, the arrangement configured ascircuit selected from the group consisting of a forward DC-DC converter;a push-pull DC-DC converter; a half-bridge DC-DC converter; adiagonal-half bridge DC-DC converter; a fall bridge DC-DC converter, acapacitor arranged for connection in parallel with the motor to limit aresulting voltage over the motor; means for measuring a current throughthe motor; and means for controlling operation of the arrangementdependent upon the measured current in the motor.

In accordance with still another aspect of the invention, there isprovided a system for driving a direct-current (DC) motor underconditions of a controlled average current, a voltage of a DC powersupply having a larger or smaller value than a motor nominal voltage,the system comprising a diode; a magnetic transformer connected inseries with the diode in a circuit arrangement selected from the groupconsisting of a flyback DC-DC converter and a ringing choke DC-DCconverter, the transformer and the diode for connection in series withthe DC motor, a switch coupled to the magnetic transformer and the diodefor connecting and disconnecting a terminal of the magnetic transformerand the diode remote from the motor to a voltage source; a capacitorarranged for connection in parallel with the motor to limit a resultingvoltage over the motor; means for measuring a current through the motor;and means for controlling operation of the switch dependent upon themeasured current in the motor.

In accordance with another aspect of the invention, there is provided asystem for driving a direct-current (DC) motor under conditions of acontrolled average current, a voltage of a DC power supply having alarger or smaller value than a motor nominal voltage, the systemcomprising an electronic synchronous rectification switch; a magnetictransformer connected in series with the synchronous rectificationswitch in a circuit arrangement selected from the group consisting of aflyback DC-DC converter and a ringing choke DC-DC converter, thetransformer and the synchronous rectification switch for connection inseries with the DC motor; a switch coupled to the magnetic transformerand the synchronous rectification switch for connecting anddisconnecting a terminal of the magnetic transformer and the synchronousrectification switch remote from the motor to a voltage source; acapacitor arranged for connection in parallel with the motor to limit aresulting voltage over the motor; means for measuring a current throughthe motor; and means for controlling operation of the switch dependentupon the measured current in the motor.

In accordance with yet another aspect of the invention, there isprovided an airflow apparatus, comprising a brush-less DC motor; anelectronic circuit for controlling operation of the brush-less DC motor;a power supply for the electronic circuit separate from a power supplyfor the brush-less DC motor, the power supply for the electronic circuitadapted to use a voltage resulting from the brush-less DC motor inoperation once the resulting voltage reaches a suitable value; and meansfor reducing power to the electronic circuit from the power supply oncethe resulting voltages reaches the suitable value.

In accordance with a further aspect of the invention, there is provideda system for powering a microprocessor based system from a DC voltagehigher than the voltage required by the system to operate, comprising acapacitor; means to charge the capacitor from the DC voltage with acurrent substantially smaller than the current the microprocessor basedsystem needs to operate; a switch coupled to the capacitor so that theswitch can connect power to the microprocessor based system from thecharge accumulated in the capacitor; means for sensing the voltage inthe capacitor and causing the switch to close once the voltage in thecapacitor reaches a desired value; and means for keeping the switchclosed while the voltage in the capacitor is over a desired value, butless than the value that caused the sensing means to close the switch.

In accordance with another aspect of the invention, there is provided aswitching based alternating current (AC) to direct current (DC)converter, comprising a rectifier adapted to be connected to analternating current (AC) mains line; a first capacitor for noisereduction connected in parallel with an output of the rectifier; aninductive element connected to a terminal of the rectifier and the firstcapacitor; a first switch coupled to the inductive element forconnecting and disconnecting a terminal of the inductive element remotefrom the parallel combination of the rectifier output and the firstcapacitor; a second switch connected to the connection node between theinductive element and the first switch, controlled so that the currentcirculating through the inductive element circulates through the secondswitch when the first switch disconnects the inductive element from theparallel combination of the rectifier output and the first capacitor; asecond capacitor for energy storage connected to the terminal of thesecond switch remote from the inductive element, the second capacitor isconnected in parallel with the serial combination of the second switchand the inductive element, the direct current (DC) output of thealternating current (AC) to direct current (DC) converter being takenfrom the terminals of the second capacitor; means for sensing a currentthrough the inductive element; means for sensing the voltage across thefirst capacitor, means for sensing the voltage across the secondcapacitor; and a control circuit connected to the first switch tomaintain the voltage across the second capacitor between defined limitsby operating the first switch in a way that a current through theinductive element tracks the waveform of the alternating current linevoltage to cause the AC to DC converter to exhibit unity power factor tothe alternating current line.

BRIEF DESCRIPTION OF THE DRAWINGS

A small number of embodiments are described with reference to thedrawings, in which:

FIG. 1 is a block diagram of an air flow generator with a common powersupply for the control electronics and the brush-less DC motor driver;

FIG. 2 is a block diagram of a brush-less DC motor included for the sakeof clarity;

FIG. 3 is a block diagram of an air flow generator with split powersupply in accordance with an embodiment of the invention;

FIG. 4 is a block diagram of an air flow generator with split powersupply and the power supply for the control electronics circuit, adaptedto use a voltage resulting from the brush-less DC motor in operationonce the resulting voltage reaches a suitable value;

FIG. 5 is a block diagram of a known DC motor driven with controlledconstant average current;

FIG. 6 is a block diagram of a system for driving a direct-current (DC)motor under conditions of controlled average current, from a voltagesource larger than a voltage that the DC motor has under operatingconditions;

FIG. 6A is a set of timing charts of the waveforms present in FIG. 6;

FIG. 6B is a more detailed block diagram of the system disclosed in FIG.6, including timing of the waveforms of one embodiment of the invention;

FIGS. 6C-1 and 6C-2 are lists of the equations that describe thebehaviour of the system shown in FIG. 6;

FIGS. 7A to 7L are block diagrams of further embodiments of theinvention relating to system depicted in FIG. 6;

FIGS. 8A to 8C are block diagrams contrasting embodiments of theinvention and the known system using switched mode power supplies;

FIGS. 9A to 9L are circuit diagrams for detailed implementations of flowgenerators in accordance with an embodiment of the invention;

FIG. 10A is a more detailed conceptual block diagram of air flowgenerator with an auxiliary power supply and an electronic switch or acurrent summing point to use a voltage resulting from the voltage on thebrush-less DC motor while in operation, once the resulting voltagereaches a suitable value;

FIG. 10B is a block diagram of a system with an auxiliary power supplyand the electronic switch;

FIG. 10C is a circuit diagram of a DC-DC converter system to adapt thevoltage developed over the motor to power the controlling electronics inaccordance with FIG. 10A;

FIG. 10D is a block diagram of a system that allows the use of a simpleauxiliary power supply without the need for low power electronics;

FIG. 10B is a circuit diagram of an electronic switch used in FIG. 10D;

FIGS. 11A and 11B are diagrams of an airflow generator for CPAPtreatment implementing an embodiment of the invention;

FIG. 12 is the block diagram of system that shows an embodiment of theinvention, in the form of a multiple cooling fan controller;

FIG. 13 is a block diagram of a system that shows another embodiment ofthe invention, in the form of a single cooling fan controller;

FIG. 14 is a block diagram of another embodiment of the invention;

FIG. 15 is a block diagram of the embodiment of the invention shown inFIG. 7J with accompanying analysis;

FIG. 16 is a block diagram of a topology for a power factor correctedswitch-mode AC to DC converter; and

FIGS. 17A to 17F are diagrams of embodiments of the invention shown inFIG. 16.

DETAILED DESCRIPTION

Systems for driving a direct-current (DC) motor under conditions of acontrolled average current, airflow apparatuses, and systems forpowering a microprocessor based system from a DC voltage higher than thevoltage required by the system to operate are described. Also describedis a typology for power factor correction. Embodiments of the inventionprovide a simple, less costly system that is obtained if the motor ismade to work as directly as possible from the AC main while powering thecontrol electronics from a smaller, simpler well regulated power supply.However, the BLDCM's available to the manufacturers of CPAP flowgenerators do not operate from the voltage available after therectification of the AC main. Further simplification, integration andversatility can be obtained if the same control electronics can lookafter the control and mode of operation of the power supply itself, andthe implementation of the functionality of the CPAP flow generator.

In other embodiments of the invention, a microprocessor can provide theintelligence necessary for controlling the functionality of the CPAPflow generator, to also control the supply of power to the BLDCMcontroller, while still providing extra functionality like power factorcorrection, and modulation of the switching frequencies used in thepower supply in order to facilitate meeting the EMI directives.

The advantage of powering the motor as directly as possible from the ACmain brings one design for a portable air flow generator for CPAPtreatment, that in terms of simplicity, uses a BLDCM that is connecteddirectly to a pulsating full wave rectified AC main. The concept isshown in FIG. 3. The control electronics (33) in FIG. 3 are powered fromthe power supply (32) that takes the power from the AC mains through afilter (31). The brush-less DC motor, BLDCM (35) that drives the turbine(36) is controlled or driven from the control electronics (34) thattakes power from the AC mains directly through the filter (31). Theelectronic commutator of the BLDCM is included in (34) while the motorsensors, that should connect to (34) are not shown for sake ofsimplicity.

As all the BLDCMs (35) suitable to build a portable air flow generatorfor CPAP in the market would not stand the main voltage directly appliedto its windings, alternative methods for driving the BLDCM must be used.

The unavailability of brush-less DC motors suitable for use in CPAP flowgenerators that can work at rectified AC main levels that has motivatedthe embodiments of the invention, which can be seen conceptually inFIGS. 4, 6, and 7.

While the embodiments of the invention are directed to motors in flowgenerators, it will be appreciated by those skilled in the art, that theembodiments of the invention are not limited to such applications.Another application of the invention is BLDCM driven cooling fans, forexample.

The power supply that the motor and the electronic commutator requirehas far more relaxed specifications than the power supply for thecontrol electronics. A motor is a forgiving load for a power supply. Themechanical characteristics of the motor work as a low pass filter, sothe motor can tolerate a relatively large ripple voltage. In fact, theripple can be up to 100% without affecting operation. If the controlledvariable is the motor current, and the electronic commutator is notpulse-width modulated, the voltage over the motor's electroniccommutator is the result of the back-emf (electromotive force) generatedin the windings themselves as a result of the motion. This voltage isapproximately a linear function of the motor speed and can be used toestimate the speed of the motor.

There are several advantages in controlling the current through a DCmotor rather than the voltage over that motor. For instance, it is wellknown that by controlling the current driving a DC motor, rather thanthe voltage over the windings, the electrical time constant formed bythe resistance and the inductance of the windings is not part of thecontrol loop dynamics, making the system more responsive. Also, the samecircuitry that is used to control the current can be used simultaneouslyto detect malfunctions due to excess current, yielding a simpler,cheaper circuit. Finally, if the current through the DC motor iscontrolled, conditions of blocked rotor are easier to handle, as thereis no excess current taken from the power source.

FIG. 5 shows a circuit for driving a DC motor (including a BLDCM withits electronic commutator) under conditions of a controlled averagecurrent.

The switch (52) closes and opens as commanded by the pulse widthmodulator (56). When the switch (52) is closed, DC voltage from thesource (51) is applied to the motor (54), a current flows through theinductance (the motor windings) embedded in the motor (L_(M)). When theswitch opens, the current flows through the diode (53). The diode (53)is usually called a “freewheeling diode” or a “catch diode”. A currentsensor (55), typically a resistor of small value, is used to measure thecurrent. Block (57) represents either an error amplifier, or acomparator, depending on the type of control system used. Block (57)compares the measured current with the desired or set current passingthe information, as control signal, to a control system (58), that inturn controls the switch (52).

This type of known system and its variations are not only applied to DCmotors but to other types of electromechanical devices with an embeddedinductance. Examples of these devices are other type of motors, likestepper motors and even solenoids.

The necessary semiconductor circuitry to perform the functions in FIG. 5can be found in integrated circuits well known to those skilled in theart (e.g., see references 5 and 19).

The details of the implementation of FIG. 5 can be found in:

-   -   reference 1, chapter 7 pages 155 to 181.    -   reference 3, chapter 11 pages 286 to 308.    -   reference 4, FIGS. 7 and 8 (technique used with other inductive        devices).    -   references 5, 6 and 13 to 19.    -   reference 24, page 3-51 to 3-70 and 5-6 to 5-9.

A limitation of the current state of the art as shown in FIG. 5 is thatwhen the switch (52) closes, the fall value of the DC voltage of thesource (51) is applied to the motor (54). Hence, the motor (54) must bespecified accordingly. A motor designed to be driven from a nominalvoltage of 24 volts DC is not, generally, expected to operate from the337 volts DC that results from rectifying a 240 VAC main.

A method for driving a motor with controlled constant average currentthrough the motor windings, while taking the power from a voltage sourcelarger than the maximum voltage source possible for the motor isdisclosed in FIG. 6.

In the circuit disclosed in FIG. 6, a switch (62) connects the DCvoltage from the power source (61) not to the motor (66) directly, butto an inductive element (64) not present in the circuit shown in FIG. 5.A capacitor (65) is added in parallel with the motor (66).

Referring to FIG. 6, the electronics controlling the operation of thecircuit (68) command a pulse width modulator (69). The object of thecontrol system (68) is to keep the current though the motor (66) asclose as possible to the desired motor current command received fromoutside the system. The control system (68) receives the informationfrom the current sensor (67) and the desired motor current as commandedfrom outside the system, and applies control algorithms or signalprocessing to obtain an output that commands the pulse width modulator(69). The operation of the control system (68) can also be described interms of a comparison of the current through the sensor (67) with thedesired motor current, as commanded from outside the system, and thegeneration of control signals for the pulse width modulator (69) inaccordance with the results of the comparison in order to achieve theobject described above.

The pulse width modulator (69) is coupled to the switch (62) and canmake the switch open and close. The switch (62) opens and closesrepeatedly, as part of a pulse width modulation scheme generated in thepulse width modulator (69) to obtain the desired current in the motor(66). Every time the switch (62) is closed, the current in the inductiveelement (64) grows in proportion to the voltage difference appliedacross the inductive element (64) and the amount of time the switch isclosed.

When the switch is open, the inductive element (64) keeps the currentflowing through the motor (66) and the capacitor (65), as the circuitcloses through the diode (63). The diode (63) is performing a similarfunction to what is normally referred to as a catch diode or afreewheeling diode. Current sensor (67) senses a current similar to thecurrent in the inductive element (64).

Generally, the capacitor (65) limits the resulting voltage over themotor, if the resulting voltage exceeds specifications for the motor dueto a relationship between the impedance of the inductive element and theimpedance of the motor. However, even if the voltage across the motorresulting from the inductive divider formed by the inductance of themotor and the inductance (64) is within the specification of the motor,the capacitor (65) is needed to take the current that the inductance(64) is forcing into the motor, during the relatively small period oftime in which the electronic commutator of the motor in a brushless DCmotor (or sometimes, the brushes in a conventional DC motor) may beopen.

FIG. 6B shows a detailed block diagram of one possible implementationthe control system (68) and the pulse width modulator (69). The controlsystem and pulse width modulator shown in FIG. 6B are a popular choicewith integrated circuits. The motor (66) of FIG. 6 has been replacedgraphically by its electrical model, comprising the inductance of thewindings (L_(M)), the resistance of the commutator and the windingR_(M), and the counter electromotive force Vemf, represented as avoltage source in series with L_(M). All the elements of FIG. 6B, withnumbers similar to the numbers used in FIG. 6, have similarfunctionality.

The capacitor (65) is shown as an ideal capacitor (65B) and itseffective series resistance (ESR), shown as a resistor labelled R_(ESR),in series with (65B).

A comparator (6B1) compares the current information in the form of avoltage from the sensing resistor (67), that is the voltage V_(RS), withthe set point, or desired motor current information, the voltageV_(DMC).

The cycle of the pulse width modulator starts periodically, when a pulsefrom the oscillator (6B3) sets the set-reset latch (6B2), causing theoutput “Q” to go high. The output “Q” of the set-reset latch (6B2) isconnected to the switch (62). The output “Q” is labelled SWC, fromswitch control. A high in SWC causes the switch (62) to close. Currentflows through the inductance (64), the capacitor (65), the motor (66)and the sensing resistor (67). When the current sensed in (67) is largerthan the set point value (V_(DMC)), the output of the comparator (6B1)is high, and the set-reset latch (6B2) is reset.

With the reset input active, the output “Q” (also labelled S_(WC))changes into low, and the switch (62) opens. The higher the value of thesignal V_(DMC), the higher the current is allowed to grow in theinductance (64), and the higher the average current through the motoris. Also, the time the switch is on (T_(ON)) grows.

The five chart timing diagram at the bottom of FIG. 6B shows the timingof some waveforms in the circuit: OSCOUT: Output of the oscillator(6B3). T is the period of the PWM signal. SWC: Control input to theswitch (62), output “Q” of the S-R latch (6B2). V_(RS): Voltage as aresult of the current flow in the sensing resistor (67). I_(L): Currentthrough the inductance (64). RESET: The output of comparator (6B1), andthe reset input of the S-R latch (6B2).

The type of control system and pulse width modulator shown in FIG. 6B isonly one example. The control schemes used with the circuit shown inFIG. 5 can be used in the control system (68) and the pulse widthmodulator (69) of FIG. 6. In fact, the circuit disclosed in FIG. 6 canbe implemented with some of integrated circuits that are currently usedto implement the circuit disclosed in FIG. 5. An example of this, is inthe reference 13.

Reference 13 uses an integrated circuit (UC3842) belonging to the samefamily (similar type of control system and pulse width modulator) thatthe integrated circuit (UC2845) used in one of the embodiments of theinvention depicted in FIG. 9H. Reference 23 is the data sheet for theUC3842.

This means that the pulse width modulation scheme used can be any of thetypes currently used to control the current through relay coils of motorwindings, when the available voltage is compatible with the maximumoperating voltage through the windings. It must be noted, however, thatwith the circuit shown in FIG. 5, there is no need for a capacitor inparallel with the motor and theta is also no need for any inductanceexternally connected in series with the motor winding.

The chosen value for capacitor (65) is not critical but cannot be toosmall. If at the frequency of operation of the pulse width modulator(69) the impedance of the capacitor (65) is negligible in comparisonwith the impedance of the motor (66), the value of the current throughthe inductive element does not depend on the characteristics of themotor (66).

Capacitor (65) may be chosen large enough to render the system formed bythe capacitor (65) coupled with the impedance of the motor and theinductance (64) at least slightly over-damped. However, the resultingcapacitor can be big. It is not necessary, if the rate of change of thecontrolled average current is limited.

Because of the current feedback system used in FIG. 6B to keep thecurrent through the motor (66) constant, independently of the motorvoltage, the inductive element (64) does not resonate with capacitor(65). Consequently, any ringing in the voltage between the motor (66)and the common terminal has a frequency (f_(Mring)) given by:f _(Mring)=1/(2Π√(L _(M) *C))where L_(M) is the inductance of the windings of the motor (see FIG.6B).

Also, if R_(RSR) is assumed to be negligible in comparison with R_(M),the condition to choose the value of capacitor (65), C, for the outputcircuit in FIG. 6B to be over-damped is:C>L _(M)*4/R _(M) ².

Capacitor (65) has to have a low ESR and good pulse characteristics. Thetype of capacitors used as filters in switched mode power supplies are asuitable choice.

In some of the embodiments of the invention, the capacitor in parallelwith the motor is also used to accumulate charge, making it possible tohave different current through the inductive element and the motor.

Sometimes, with the known systems (FIG. 5), a small value capacitor,typically ceramic high frequency capacitor, is used to filtercommutation noise in DC motors to improve EMI compliance. This capacitorcannot be confused with capacitor (65).

If the noise filtering capacitor of the current state of the art isremoved, the motor continues to operate. However, this only happens withthe embodiment of the invention, if the inductance of the motor has avalue such that the inductive divider formed with the inductive element(64) yields a resulting voltage compatible with the safe operation ofthe motor. With the capacitor (65) in the circuit, the impedance of themotor conducts only DC current in steady state, and hence is not part ofthe dynamic of the system controlling the current. With some systems, itwould be an advantage not to have current at the frequency of operationof the PWM (69) traveling through the wires to connect to the motor(66), especially if the frequency is high to have a physically smallinductive element (64).

Some of the pulse-width modulation schemes known that can be used forthe circuit disclosed in FIG. 6, are listed below:

-   -   1. Hysteresis PWM    -   Cycle by cycle current control. A hysteresis band is set around        the desired current level. The top of the hysteresis band        determines when the switch is turned on and conversely the        bottom of the band determines when to turn the switch off.        Ripple current is determined by the difference between the        levels and the speed of reaction of the circuitry. Must be able        to sense the current during the whole cycle. Frequency is        determined by the value of the currents, the value of the DC        voltage source and the value of the energy (current) storing        inductance.    -   2. Clocked turn-ON, constant frequency PWM    -   Cycle by cycle current control. A periodic signal (clock) starts        the cycle by turning the switch on. At a desired current level        the switch is turned off. The frequency is determined by the        frequency of the periodic signal. It is necessary to measure the        current while the switch is on, and not during the whole cycle.        It is the most commonly used PWM in systems with current mode        control. It is the system shown in FIG. 6B, used in the IC        chosen for the detailed implementation shown in FIG. 9H (UC2845)        and implemented with small scale integrated circuits in FIG.        11B.    -   3. Clocked turn-OFF, constant frequency PWM    -   Cycle by cycle current control. Like the type 2 above, but the        periodic signal turns the switch off, to be turned on when the        current has dropped under a set value. It is not necessary to        measure the current during the whole cycle. The frequency is        determined by the frequency of the periodic signal.    -   4. Two current levels PWM (also called triangle PWM)    -   Cycle by cycle current control. One current level to turn the        switch on, another to turn the switch off. Ripple current        determined by the difference between the levels and the speed of        reaction of the circuitry. Must be able to sense the current        during the whole cyclo. Frequency is determined by the value of        the currents, the DC voltage source and the value of the energy        (current) storing inductance.    -   5. Constant OFF time PWM    -   Cycle by cycle or average current control. The cycle starts by        turning the switch on. When the current reaches a pre-set level,        the switch is turned off and stays off for a fixed amount of        time, to complete the cycle. At the end of the fixed period of        time, the switch is turned on again, starting a new cycle. It is        necessary to measure the current while the switch is on, and not        during the whole cycle. Frequency is determined by the value of        the current, the value of the DC voltage source, the value of        the energy (current) storing inductance and the fixed time off        of the switch. If the average current over a number of cycles is        controlled, the on time is made to vary as a result of the        difference between the desired current and the actual current.    -   6. Constant ON time, average current PWM    -   The switch is closed for a fixed period of time. At the end of        the fixed period, there is a variable delay, and the cycle        starts again. The frequency is variable. The average current        over a number of cycles is controlled, rather than during each        cycle. The off time is made to vary as a result of the        difference between the desired current and the actual current.    -   7. Constant frequency, variable duty cycle PWM    -   The duty cycle of a square wave is made to vary as a result of        the difference between the desired current and the actual        current. The average current over a number of cycles is        controlled, rather than during each cycle. It is the most widely        used choice in voltage control mode switch mode power supply        integrated circuits. It is also available as a peripheral in        microcontrollers.    -   8. All variable (On time, Off time and the Frequency)    -   This method is described in reference 7 at page 64.

A more detailed description of the operation of the pulse-widthmodulation schemes above can be found in:

-   -   Reference 1 pages 174 to 179    -   Reference 3 pages 64, 93, 104 to 145, 249 and 297    -   Reference 12 pages 70 to 75

FIG. 6A shows the idealised waveforms in the system depicted in FIG. 6.The names in the charts are referencing points in the embodiment of FIG.6B, however, as only two cycles are shown, any of the above listed pulsewidth modulations schemes give similar waveforms. Obviously, if enoughamount of cycles would be considered, different pulse width modulatorsshow different timing, since for instance, some change the frequency ofoperation. However, there always is a “T_(ON)” and a “T_(OFF)”, theanalysis can be made in a similar way, the results will be similar, andwith all of them, a similar value of average current can be obtained.

In FIG. 6A, the waveform in the charts, are: V_(A): Voltage betweenpoint A in FIG. 6B and ground. V_(FD): Forward voltage diode (63).V_(S): Voltage across the switch (62). V_(DSW): Voltage-drop across theswitch (62) when the current I_(L) flows. V_(L): Voltage across theinductive element (64). V_(LP): Peak value of V_(L). V_(LM): Minimumvalue of V_(L). I_(L): Instantaneous current across the inductiveelement (64). Δ_(IL): Difference between the maximum and the minimumvalues of I_(L). I_(LPK): Peak value of I_(L). I_(AVERAGE): Averagevalue of I_(L). Is similar to the current through the motor I_(M). T:Period of the pulse width modulator (It can be variable). I_(M): Motorcurrent, the variable that is controlled, by changing the PWM. I_(C):Current through the capacitor (65). T_(ON): Period of time the switch(62) is closed, SWC must be high. T_(OFF): Period of time the switch(62) is open, SWC must be low. V_(C): Voltage over (C65B) the capacitivepart of the model of the real capacitor (65). V_(ESR): Voltage overR_(ESR) the resistive part of the model of real capacitor (65). V_(M):Voltage as indicated in FIG. (6B). SWC: Control input to the switch(62). A high value causes the (62) to close. I_(S): Current through theswitch (62). I_(D): Current though the diode (63).

For all the charts, the independent variable is the time.

Some of the charts in FIG. 6A have the dependent variable not in scale.This was thought to be a better choice than letting the small value ofsome variables like V_(C), V_(DSW) and V_(FD), hide their existence inthe charts. If the power source (61) in FIG. 6 is rectified AC mainvoltage, the value of V_(LP), in the chart labelled V_(L), has to bedrawn out of scale, as shown in the chart.

FIGS. 6C-1 and 6C-2 have the equations describing the operation of thecircuit disclosed in FIG. 6, using the embodiment shown in FIG. 6B. Theequations hold true even if a different pulse width modulation scheme isused. With some types of pulse width modulator, the frequency is notconstant. The equations, however, are correct for each value of theparameters, once steady state is reached. Most of the variables havealready been listed above, the exceptions are: V_(DC): Value of thevoltage of the DC power source (61). R_(M): The resistance of thewindings and commutator of the motor (66). Vemf: Counter electromotiveforce of motor (66). It is the voltage induced in the windings as themotor axis rotates. Kv: DC motor constant. The voltage of Vemf inducedfor unit of angular speed (Volt * sec). ω: Angular speed of motor axis(1/sec). V_(RS): Voltage that is the result of the current been measureusing the sensing resistor (67). R_(S): Value of the sensing resistorused to measure the current. The resistor value has to be negligible incomparison with the resistance of the motor R_(M). R_(ESR): Effectiveserics resistance of the real capacitor (65). I_(LMIN): Minimum value ofthe current through the inductance (64). F: Frequency of the pulse widthmodulator F = 1/T.

The analysis used the fact that as the impedance of the capacitor (65)at the frequency of operation F is negligible in comparison with theimpedance of the motor (66), all the AC components (the terms containinga multiple of the angular frequency, in the Fourier series expansion) ofthe pulse-width modulated waveform can be considered to circulatethrough the capacitor. That sum of terms is what makes the currentI_(C). The rest of the current through the inductive element (64)circulates through the motor (66). The current through the realcapacitor (65) causes a voltage drop, due to the impedance of thecapacitive (C) component of the capacitor (65) and also its effectiveseries resistance R_(ESR). For the analysis to be accurate, this voltagedrop has to be negligible in comparison with the motor voltage V_(M).

Equation (3) in FIG. 6C-1 shows this assumption, as it states that themotor voltage is approximately equal to I_(M)*R_(M)+Vemf.

The equations in FIG. 6C-1 result from applying Kirchhoff's laws to FIG.6B.

It is also assumed that the inductive element (64) has no seriesresistor associated with the inductive element. Hence the equation (15)is true and results in equations (16), (17) and (18), of FIG. 6C-1, whenthe switch (62) is closed, and in equation (19), of FIG. 6C-2, when theswitch is open and the current is circulating through the diode (63).

Equation (20) shows, as it can be seen in the chart I_(L) of FIG. 6A,that in steady state, the net change of current through the inductiveelement per cycle is zero. That is, the current grows when the switch isclosed and is equal to the current drop when the switch is open.

Equation (21) eliminates the voltage drop over the switch (62) (V_(S))and the forward voltage drop over the diode (63) (V_(FD)), as they arenegligible in comparison with the terms they are subtracted from oradded to.

Equation (22), follows from equation (21) and gives the relationshipthat the duty cycle (T_(ON)/T) of the pulse width modulator has with themotor voltage, as a consequence of the motor current.

This system differs from a current driven buck converter in that thevoltage over the load is not a regulated variable, but it is the resultof the effect of the current itself in the load. In a DC motor, thevoltage in the load is related to the back electromotive force (Vemf)induced in the motor windings as a result of the motion of the windingsin the magnetic fields inside the motor. In fact, given the typicalvalues of resistor RM, and the voltage drop caused on it by the motorcurrent, VM can be used as an estimate of the speed of the motor.

The differences with the current fed buck converter and the well knowndown converter switched mode power supply are: DF_1. Under normaloperation, the voltage in the load is not taken into account by thecontrol loop, while in a switched mode power supply, is the controlledvariable. With a DC motor, its voltage depends on the back emf. Hence,the voltage depends on the speed of rotation of the motor axis. DF_2. Ina switched mode power supply, the capacitor in parallel with the loadfilters the ripple voltage and may provide the current to the load whenthe energy accumulated in the inductance drops to zero before the end ofthe period of the pulse width modulator (what is normally calleddiscontinuous operation). In the embodiment of the invention however, ifthe current through the inductance drops to zero, the inertia of themotor keeps the motor rotating, and the voltage over the motor is keptby the electromotive force induced in the winding as the motor axisrotates.

The analysis of the system of FIG. 6B continues in FIG. 6C-2 for the ACcomponent of the voltage in V_(M) due to the voltage drop over thecapacitor (65) while in steady state.

The graph in FIG. 6C-2 is a detailed enlargement of the chart labelledI_(C) of FIG. 6A. It has to be examined keeping in mind that the currentthrough capacitor (65) is similar to the current through the inductiveelement (64), where the average current I_(AVERAGE) has been subtractedfrom. The area shaded in the drawing, labelled ΔQ, is the total amountof charge that is transported during one half of the cycle of the pulsewidth modulator, in or out of the capacitor (65), from the inductiveelement (64).

Equation (23) in FIG. 6C-2 is the area of the triangle that has beenshaded in the drawing (half the base multiply by the height). Equation24 in FIG. 6C-2 shows the relationship between the current variationsthrough the inductive element (64) and the peak voltage in thecapacitive component (C) of the model of capacitor (65).

To get the voltage over capacitor (65), the voltage drop over theresistive component R_(ESR) of the model of capacitor (65) must beadded, as is shown in equation (25) of FIG. 6C-2.

If the typical figures for the variables in equation (25) are replaced,the result of equation (26) of FIG. 6C-2 is reached, validating theoriginal assumption regarding V_(M) in equation (3) of FIG. 6-1.

Numerous variations in the implementation of the circuit disclosed canbe made.

The inductance, for instance, can be the inductance of the winding of atransformer. In this case, other windings could produce uncontrolledvoltages that can be used to power electronic circuits. This scheme hasbeen widely used in switched mode power supplies.

Another modification of the system disclosed above is that the sense ofthe current can be changed by changing the polarity of the voltagesource. This can be easily done by replacing the switch in FIG. 6 with abridge circuit as is commonly done in four-quadrant control of DCmotors. Also the embodiment of the invention can be applied to what isknown as universal motors, a motor with conventional brushes that isinsensitive to the sense of the current through it, and it is normallydriven by an AC voltage source.

In yet another modification, the capacitor (65) in FIG. 6 can bepresent, not as an individual component, but embedded in other circuitblocks, as for instance, the electronic commutator of a brushless DCmotor if the motor (66) is a brushless DC motor, or other controlelectronics coupled with the motor (66).

All the strategies of pulse-width modulation used with the known systemscan be used with the embodiment of the invention to implement thecontrol system (68) and the pulse-width modulator (69). Similarly, theswitch (62), the diode (63) and the current sensor (67) have the samefunction as with the current state of the art and can be designedfollowing criteria well known to those skilled in the art.

Some of the numerous possible variations of the circuit shown in FIG. 6are shown in FIGS. 7A to FIG. 7L.

FIG. 7A shows a system similar to FIG. 6 with a different topology. Thepower source (7A01) has a similar function to the voltage source (61) inFIG. 6. Motor (7A02) has a similar function to the motor (66) in FIG. 6.Capacitor (7A03) has a similar function to the capacitor (65) in FIG. 6.Diode (7A04) has a similar function to the diode (63) in FIG. 6. Theinductive element (7A05) has a similar function to the inductance (64)in FIG. 6. Control system (7A07) has a similar function to the controlsystem (68) in FIG. 6. Pulse width modulator (7A06) has a similarfunction to the modulator (69) in FIG. 6. Current sensor (7A09) has asimilar function to the current sensor (67) of FIG. 6. The switch (7A08)has a similar function to the switch (62) of FIG. 6, however, switch(7A08) is connected to ground rather than the positive terminal of thepower source (7A01). This circuit makes the implementation of theelectronics switch (7A08) and its drivers easier (or simpler) than theelectronic switch (62) in FIG. 6. However, the motor (7A02) is removedfrom the common terminal of the circuit.

Although electrically it appears to be similar to swap the position ofthe inductive element (7A05) and the parallel combination of the motor(7A02) and the capacitor (7A03), in practice, it is better to connect asshown in FIG. 7A. The reason for this is that in the way shown, theisolation of control elements connected to circuits in the motor aresubject to less AC voltage stress with respect to grounded circuitblocks.

A detailed implementation of this topology is shown in FIGS. 9A to 9L.

FIG. 7B shows that there are many points to measure the current for thecontrol system of the circuit disclosed in FIG. 6. FIG. 7B follows thesame architecture than FIG. 6. In this figure, the blocks (68) and (69)of FIG. 6 have been joined together in a single control system (7B09).

(7B01) is similar to (61). (7B10) is similar to (62). (7B06) is similarto (64). (7B11) is similar to (65). (7B12) is similar to (66). (7B7) issimilar to (63).

All the components 7B02, 7B03, 7B04, 7B05 and 7B06 are current sensors.However, as will be apparent to those skilled in the art, not all thepositions render the same waveform. Furthermore, if the sensor is notsensing current in a branch of the circuit connected to ground (e.g., in(7B02) and (7B06), differential amplification techniques like thecircuit shown on (7B14) must be used. The current in (7B03), (7B05) and(7B04) can be sensed with a simple small value resistor withoutdifferential techniques. A current transformer like the one illustratedin (7B13) can be used in all the locations. Also, the current in (7B02)flows when the current in (7B03) is not flowing. Current in (7B06) issimilar to (7B04). With the values normally chosen for the capacitor(7B11), at the frequency of operation normally chosen to control theswitch (7B10), all the high frequency components of the current flowthrough (7B11) and the low frequency component flow through the sensorin (7B05).

Hence a sensor in (7B05) cannot be used to control the current on acycle-by-cycle basis, but only as the average over a certain number ofcycles of the pulse width modulator embedded in (7B09). Reference 10,chapters 13 and 14, pages 3.172 to 3.192 provide useful information forimplementation of current sensors.

FIG. 7C shows a system similar in topology to FIG. 6. In this system,the pulse-width modulator (69) and the control system (68) of FIG. 6 areincluded in the control system 7C08. If it is compared with FIG. 6above: (7C01) is similar to (61); (7C02) is similar to (62); (7C04) issimilar to (64); (7C06) is similar to (65); (7C05) is similar to (66),and current sensor (7C07) is similar to (67).

The difference with FIG. 6 is a switch (7C03) in the position of thediode the other figures (63, 7A04 and 7B07). Control system (7C08) hasan extra output to control the additional switch (7C03). Those skilledin the art of designing switched mode power supplies will recognise thetechnique known as “synchronous rectification”. This technique consistsof replacing the diodes by switches that dissipate less power than thediodes. The switches have to be closed or opened at the precise momentthat the diode that they replace conducts (switch close) or is blocking(switch opens). This technique is currently popular when the current ishigh, to increase the efficiently of switched mode power supplies,especially at low output voltages. In practice, as the power MOSFETtransistors normally used to implement the switches also include a diodein parallel that can conduct when the switch performing the synchronousrectification is not yet closed, the synchronization does not need to beperfect. The control signals to the switches are the logical complementof each other. The controller (7C08) must not close both switches (7C02)and (7C03) at the same time or the power source (7C01) is shorted toground. Each time the switches must change their state, a period of timein which both switches are commanded to be open must be introduced. Thistime is what is normally known as “dead time”.

In the case of the switch replacing the diode (7C03), the diode inparallel with the switch starts conducting before the switch is made toclose. This causes a negligible loss in efficiency. Switch (7C03) canhave an actual diode in parallel in addition to the diode embedded inthe MOSFET transistor, to facilitate the implementation.

Reference 12, page 60 gives details of synchronous rectification. Also,reference 24, section 2.2.3, pages 2-76 “An introduction to synchronousrectifier circuits using PowerMOS transistors” and reference 2, providean implementation of synchronous rectification.

FIG. 7D shows a generalisation of the circuit shown so far. The pulsewidth modulator (69) in FIG. 6 and (7A07) in FIG. 7 and the electronicswitch (62) or (7A08), for instance, are simply providing alternativelya voltage close V_(DC) or an open circuit (close to 0 voltage, asmeasured from the cathode of the diode (63) in FIG. 6).

Referencing FIG. 6, or FIG. 6B, any technique that provides acontrollable pulse width modulated waveform in the point (A), extractingpower from the DC power source (61) to be delivered to the componentsconnected to point A, can be used with results similar to what thesystem in FIG. 6 can achieve with the switch (62).

A magnetic device, like a transformer, coupled with a plurality ofswitches and diodes can be used to that end. In view of the foregoing,it is possible to generalise and define a Power Pulse Width Modulator(PPWM) block (7D02) comprising a plurality of switches in block (7D03)coupled with a magnetic device, also coupled to a plurality of diodes,the latter combination in block (7D04).

Blocks like (7D02) are currently found in switched mode power supplies.The switches in block (7D03) inside PWM (7D02) are controlled from thecontrol system 7D09.

The control system (7D09) includes the control system (68) and the pulsewidth modulator (69). It differs from similar blocks found in FIGS. 6and 7A, in that if there is more than one switch, the system has to havemore than one control line to command it to open or close.

Not shown in FIG. 7D, for sake of clarity, is the fact that in somecombinations with more than one switch, it may be necessary to measurethe current in more than one point. In those cases, control systems like(7D09) also differ firm the combination (68) and (69) of FIG. 6, in thatthey receive current information from more than one sensor.

All other parts of the system shown in FIG. 7D are equivalent infunction to part of the system shown in FIG. 6. Specifically, withreference to FIG. 6: (7D01) is similar in functionality to the powersource (61); (7D06) is similar in functionality to the inductive element(64); (7D08) is similar in functionality to capacitor (65); (7D07) issimilar in functionality to motor (66), (7D10) is similar infunctionality to current sensor (67); and (7D05) is similar infunctionality to diode (63).

The concept behind FIG. 7D will be more clearly understood in light ofthe following figures showing topologies currently used with switchedmode power supplies applied to the circuit disclosed in FIG. 6. In eachcase, there is a different implementation of the PPWM block (7D02).

The reasons for adding a transformer to the circuit disclosed in FIG. 6are:

-   -   Possibility of isolation between the power source (7D01) and the        motor (7D07) or the control system (7D09).    -   The voltage of the power source (7D01) is not limited to values        larger than the operating voltage of the motor (7D07).

The reason for not simply using an off-the-shelf switched mode powersupply and drive the motor in current control mode using the circuitshown in FIG. 5 is that by implementing the circuit in the mannerdisclosed, a simpler, less expensive system results, especially if thesystem has to provide quick accelerations of the motor and power factorcorrection.

The current levels in the secondary side of the transformers (the motorside) are observed in the primary side (the switch or switches side)affected by the turns ratio of the transformer. This must be taken intoaccount while designing the switches, and when placing current sensorsin the primary side of the transformers.

After the generalisation made in FIG. 7D, four examples follow:

EXAMPLE 1 FIG. 7E. Topology Derived from the Forward Converter

The system in FIG. 7B uses a transformer (7E03) to couple a rectangularwaveform to the node labelled (A) in the drawing. There is a pulse widthmodulator embedded in the control system (7E11) that controls the switch(7E09) and can command it to open or close. When the switch (7E09) opensand closes, a rectangular waveform is applied to the primary side of thetransformer (7E03). This rectangular waveform is coupled by atransformer (7E03) to its secondary side and is coupled to the nodelabelled (A) in the drawing, through the diode (7E04). The node labelled(A) in the drawing is coupled to a system similar to the one found inFIG. 6. If FIG. 7B is compared with a combination of FIG. (6) and FIG.(7A), the following applies: power source (7E01) is similar infunctionality to the power source (61); winding or inductive element(7E05) is similar in functionality to the inductive element (64);capacitor (7E07) is similar in functionality to capacitor (65). Motor(7E08) is similar in functionality to motor (66), (7E10) is similar infunctionality to current sensor (7A09); (7E09) is similar infunctionality to the switch (7A08); and (7E06) is similar infunctionality to diode (63). The pulse width modulator (7A06) and thecontrol system (7A07) are embedded in the control system (7E11).

The function of diode (7E04) is to prevent the winding of transformer(7E03) short circuiting the DC voltage in the point labelled (A) in thedrawing to ground. Diode (7E02) discharge the energy accumulated in theinductance of the transformer (7E03), while the switch (7E09) is open.With this type of connection, the duty cycle of the pulse widthmodulator inside (7E11) has to be less than 0.5 (50%).

The same pulse width modulator detailed in FIG. (6B) can be used in(7E11), using the small resistor (67) of FIG. 6B to implement the sensor(7E10). However, it must be kept in mind that the current sensed in theposition of sensor (7E10) is the current though the inductive element(7E05), affected by the number turn relationship of transformer (7E03).

Clearly, the combination of (7E02), (7E03), (7E04) and (7E09) implementsthe block (7D02) of FIG. 7D.

Those skilled in the art of switched mode power supplies, rat than motorcontrol will recognise that the system of FIG. 7E has pad of thetopology of a current mode “forward converter”.

Details of how to design each of the components coupled with thetransformer (7E03) and the transformer itself can be found in thefollowing references:

-   -   Reference 3, page 217.    -   Reference 7, page 104 and 170.    -   Reference 9, page 37.    -   Reference 10, pages 2.63 to 2.69.    -   Reference 11, pages 76 to 83.    -   Reference 12, page 32, and 40.    -   Reference 24, page 2-13, 2-84 and 2-101.

EXAMPLE 2 FIG. 7F. Topology Derived from the Push-Pull Converter

FIG. 7F uses a transformer (7F04), two switches (7F02) and (7F12), andtwo diodes (7F05) and (7P06) to implement the block (7D02) in FIG. (7D).The two switches alternate in opening and closing. The combination ofelements named above, generates a pulse width modulated waveform in thenode labelled (A) in the drawing. The node labelled (A) in the drawingis coupled to a system similar to the one found in FIG. 6. If FIG. 7F iscompared with a combination of FIG. 6 and FIG. 7A, the followingapplies: (7F01) is similar in functionality to the power source (61);(7F08) is similar in functionality to the inductive element (64); (7F09)is similar in functionality to capacitor (65). (7F10) is similar infunctionality to motor (66), (7F13) and (7F03) are similar infunctionality to current sensor (7A09); (7F02) and (7F12) are similar infunctionality to the switch (7A08); and (7F07) is similar infunctionality to diode (63). The pulse width modulator (7A06) and thecontrol system (7A07) are embedded in the control system (7F11).

The function of diodes (7F05) and (7F06) is to prevent the winding oftransformer (7F04) from short circuiting the DC voltage in the pointlabeled (A) in the drawing to ground.

The control system (7F11) has the functionality of the control system(7A07) and the pulse width modulator (7A06). One difference is thatcontrol system (7F11) has to take information from two current sensors(7F03) and (7F13). However, since when one switch is open the other isclosed, the current information from both sensors can simply be addedtogether, and can be filtered with a low pass filter, to get a waveformsimilar to the waveform of sensor (67) in FIG. 6. It must be kept inmind that the current sensed by the sensors is the current through theinductive element (7F08), affected by the number of turns relationshipof the transformer (7F04). The other difference is that consistentlywith the use of two switches, and as it was mentioned while commentingon FIG. 7D, the control system (7F11) has two outputs to control oneswitch each.

The system formed by transformer (7F04), diodes (7F05) and (7F06) andthe switches (7F02) and (7F12) can be analysed and designed withtechniques similar to the techniques used in a “push-pull” switched modepower supply.

Details of how to design each of the components coupled with thetransformer (7F04), and the transformer itself can be found in thefollowing references:

-   -   Reference 3, page 220.    -   Reference 7, page 116 and 153.    -   Reference 9, page 37.    -   Reference 10, pages 2.147 to 2.151 and 2.153 to 2.159.    -   Reference 11, pages 34 to 38.

EXAMPLE 3 FIG. 7G. Topology Derived from Half and Full Bridge Converter

In FIG. 70, a block equivalent to the block (7D02) of FIG. (7D) isimplemented with capacitors (7002) and (7014), switches (7G03) and(7004), diodes (7G06) and (7G07) and transformer (7G05).

The two switches alternate in opening and closing. The combination ofelements named above, generates a pulse width modulated waveform in thenode labelled (A) in the drawing.

The node labelled (A) in the drawing is coupled to a system similar tothe one found in FIG. 6. If FIG. 7G is compared with a combination ofFIG. 6 and FIG. 7A, the following applies: (7G01) is similar infunctionality to the power source (61); (7G09) is similar infunctionality to the inductive element (64); (7G11) is similar infunctionality to capacitor (65). (7G10) is similar in functionality tomotor (66), (7G12) is similar in functionality to current sensor (67);(7G03) is similar in functionality to the switch (62) and (7G04) issimilar in functionality to the switch (7A08); and (7G08) is similar infunctionality to diode (63).

The pulse width modulator (69) and the control system (68) are embeddedin the control system (7G13).

The function of diodes (7G06) and (7G07) is to prevent the winding oftransformer (7G05) to short circuit the DC voltage in the point labelled(A) in the drawing to ground.

The control system (7G13) has the functionality of the control system(68) and the pulse width modulator (69). The current sensor 7G12 can beimplemented with the small resistor of FIG. 6B. The control system mustcontrol two switches, so as mentioned while commenting on FIG. 7D, thereare two control outputs.

Those skilled in the art of switched mode power supplies, rather thanmotor control, will recognise the transformer arrangement of the“half-bridge” switched mode power supply. Furthermore, by looking atFIG. 7G, it will be obvious to those skilled in the art of switched modepower supplies, rather than motor control, that replacing the capacitors(7G03) and (7G14) with diodes with the cathode connected towards thepositive terminal of the power source (7G01) provides a “diagonal halfbridge” (sometimes called a “two transistor forward converter”).Further, by replacing the capacitors (7G03) and (7G14) with electronicswitches controlled by adding two extra switch control lines to thecontroller in FIG. 7G 13, the “bridge”—also known as a “fall bridge”configuration is obtained.

Details of how to design each of the components coupled with thetransformer (7G05), and the transformer itself can be found in thefollowing references:

-   -   Reference 3, page 223.    -   Reference 7, pages 111, 113 and 152.    -   Reference 9, page 93.    -   Reference 10, pages 2.80 to 2.115.    -   Reference 12, pages 33.

EXAMPLE 4 FIG. 7X. Topology Derived from the Flyback Converter

This embodiment is based on the switched mode power supply topologyknown as a flyback converter (also known as “ringing choke powersupply”). In this case, the system is different from FIG. 7D, since asingle component, the transformer (7H03) provides the functionality ofboth, the inductive element (64) and the catch or freewheeling diode(63) in FIG. 6. The inductive element (64) is provided by thetransformer inductance, and the diode (63) is not needed because of theproperties of the magnetic system formed by the transformer (7H03) ifits windings are corrected with opposing polarities. Transformersworking in the mode (7H03) is working, has also been described in theliteratures as two coupled inductors.

Diode (7H04) is used simply to prevent the secondary winding of thetransformer to short circuit the positive motor voltage to ground.

Snubbers circuits in blocks (7H02) and (7H07) are needed due to the factthat a practical transformer (7H03) has parasitic components, like forinstance leakage inductance.

If FIG. 7H is compared with a combination of FIG. 6 and FIG. 7A, thefollowing applies: (7H01) is similar in functionality to the powersource (61); the functionality of the inductive element (64) is providedby transformer (7H03); (7H05) is similar in functionality to capacitor(65). (7H06) is similar in functionality to motor (66), (7H09) issimilar in functionality to current sensor (7A09); (7H08) is similar infunctionality to the switch (7A08); the functionality of diode (63) isprovided by transformer (7H03). The pulse width modulator (7A06) and thecontrol system (7A07) are embedded in the control system (7H10).

The control system (7H10) can be implemented with the pulse widthmodulation method shown in FIG. 6B. This is a frequently used method forcontrolling current through a flyback transformer. An integrated circuitoften used to implement this functionality in switched mode powersupplies is the UC2845 (see reference 23).

Details of how a flyback transformer operates, how to design each of thecomponents coupled with the transformer (7H03), and the transformeritself can be found in the following references:

-   -   Reference 3, page 214.    -   Reference 7, pages 151, 166 and 190.    -   Reference 9, page 105.    -   Reference 10, pages 2.3 to 2.62.    -   Reference 11, pages 29 to 34 and 83 to 90.    -   Reference 12, pages 32, 36 to 34 and 83 to 90.

EXAMPLE 5 FIG. 7J New Topology Derived from an Un-Insulated InputReferenced Flyback Converter or from an Input Referenced Boost Converter

This embodiment is loosely based on the switched mode power supplytopology known as a flyback converter (also known as “ringing chokepower supply”). However, it is different in the sense that it does notuse a transformer and the output is referred to the input and not toground.

The circuit works as follows: When the switch (7J06) closes, currentflows from the DC voltage source (7J01) to the inductive element (7J05).Diode (7J04) is reverse-biased. When the switch (7J06) opens, thecurrent circulating through the inductive element flows through thediode (7J04), to the parallel combination of the capacitor (7J02) andthe motor (7J03).

If FIG. 7J is compared with FIG. 7A, the following applies: (7J01) issimilar in functionality to the power source (7A01); the functionalityof the inductive element (7A05) is provided by (7J05); (7J02) is similarin functionality to capacitor (7A03); (7J03) is similar in functionalityto motor (7A02); (7J07) is similar in functionality to current sensor(7A09); (7J06) is similar in functionality to the switch (7A08); thefunctionality of diode (7A04) is provided by (7J04). The pulse widthmodulator (7A06) and the control system (7A07) are embedded in thecontrol system (7J08).

The control system (7J08) can be implemented with the pulse widthmodulation method shown in FIG. 6B.

There is currently no switch mode power supply topology using aconfiguration from which the circuit can be derived. An analysis of thecircuit operation is provided in FIG. 15, for the case of continuouscurrent flow though the inductance (7J05).

Once the analysis is done, it can be learned from it, that the dynamicsof the system can be described with equations similar to the buck-boostconverter (also known as the positive to negative converter) howevercare should be taken to keep in mind the polarity of the output and thefact that it is not referenced to ground. The analysis of the dynamicbehaviour of the buck-boost converter (also known as the positive tonegative converter) can be found in references 3, 9 and 10.

The meaning of the notation used in FIG. 15, can be seen directly fromthe circuit diagrams in FIG. 15. The analysis is made for the case ofcontinuous current circulation through the inductive element. The resultin equation (1506) shows that the voltage over the motor V_(M), can belower or higher than the value voltage source (7J01). The relationshipbetween the both voltages depends on the relationship between T_(ON) andT_(OFF).

The Equations (1501) and (1502) are similar to the equations (3) and (4)in FIG. 6C-1. Equation (1503) shows that it is assumed that the voltagedrop in the inductive element (7J05) is must larger in value than thevoltage drop in the switch (7J06) and the current sensor (7J07).Similarly equation (1504) shows that it is assumed that the voltage dropin the diode (7J04) is much smaller than the value of the voltage overthe motor.

It will be obvious to the skilled in the art how to do the same analysisfor the discontinuous case, wherein the current though the inductiveelement drops to zero before T_(OFF) finish. This type of analysis iscommon to switch mode power supply technology. In any case, the resultsfrom the analysis of the buck-boost converter (negative to positiveconverter) can be used with the precautions explained above.

EXAMPLE 6 FIG. 7L Topology Derived from the Buck-Boost Converter

This embodiment is based on the switched mode power supply topologyknown as a buck-boost converter (also known as a positive to negativeconverter). The circuit works as follows: When the switch (7L06) closes,current flows from the DC voltage source (7L01) to the inductive element(7L05). Diode (7L04) is reverse-biased. When the switch (7L06) opens,the current circulating through the inductive element will flow throughthe diode (7L04) to the parallel combination of the capacitor (7L02) andthe motor (7L03).

If FIG. 7L is compared with a combination of FIG. 6 and FIG. 7A, thefollowing applies: (7L01) is similar in functionality to the powersource (61); the functionality of the inductive element (64) is providedby (7L05); (7L02) is similar in functionality to capacitor (65). (7L03)is similar in functionality to motor (66), (7L07) is similar infunctionality to current sensor (7A09); (7L06) is similar infunctionality to the switch (62); the functionality of diode (63) isprovided by (7L04). The pulse width modulator (69) and the controlsystem (68) are embedded in the control system (7L08).

The control system (7L08) can be implemented with the pulse widthmodulation method shown in FIG. 6B.

For this embodiment there is a well-known switch-mode power supplytopology from which the steady state operation can be derived (See ref.9, page 150, ref. 10 page 31 and ref. 3 page 81).

For the case of continuous current through the inductance, the resultsare similar to the circuit 7J. Hence the voltage over the motor can belarger or smaller than the voltage of the DC power source.

As with all the other examples, the voltage over the motor for the givencurrent set by the control system (7L08), will depend on thecharacteristics of the motor, its load, and the speed of rotation.

The advantage of this circuit with respect to the circuit in FIG. 7J, isthat the motor is referred to ground, however the voltage is of negativepolarity with respect to the voltage of the DC power source.

FIG. 7I illustrates a different embodiment of the circuit disclosed inFIG. 6. The main difference is that the current through the motor (7I7)as measured with the current sensor (7I12) is controlled independentlyof the current through the inductance L_(M) (7I5), which is measuredwith the current sensor (7I11). The current through the inductance (7I5)is controlled by applying a pulse width modulated waveform to thecontrol terminal of the switch (7I3) in a way similar to FIG. 6. Thecurrent through the motor is controlled by applying pulse widthmodulation techniques to the switches in the electronic commutator (7I6)in FIG. 7I. In this case, the motor current is controlled using theinductance of the motor windings, exactly as the current state of theart does. This is shown in FIG. 5. FIG. 7I shows two different controlsystems, for sake of clarity, however, in practice, a singlemicroprocessor can control both current loops. In this case, the setcurrent signal shown from the main controller (7I10) to the controller(7I8) is a value passed between different segments of a running program.

The system of FIG. 7I can revert to the system of FIG. 7A, if theelectronic commutator (7I6) of the motor (7I7) is not pulse-widthmodulated. In this case the current through the motor 7I7 is the lowfrequency component of the current switched through the inductance L_(M)(7I5) by the switch (7I3) and the diode (7I4).

If the current through the motor (7I7) is made different than thecurrent provided through the coil (7I5), charge accumulates in capacitorC_(M) (7I9) raising the voltage of the capacitor.

Hence the current through the motor can still be controlledindependently of the current through the coil L_(M), in all the otherembodiments, without sensing the actual motor current, simply bymeasuring the voltage in the capacitor with relation to what the voltageshould be if it were the product of the back electromotive force inducedvoltage in the motor windings.

Allowing control of the current through the inductance (7I5)independently of the current provided to the system comprising the motor(7I7) its electronic commutator (7I6) and the capacitor (7I9) yields avery versatile apparatus. There are at least two applications for such asystem:

Appplication_(—)7I_(—)1.—Power Factor Correction

The capacitor in the rectifier filter (7I2) is chosen small enough forthe voltage at the output of the rectifier (7I1) to track the full-wave,rectified AC main voltage. The current in the inductance L_(M) (7I5) ismade a full wave rectified waveform:I(t)=Ipeak*|sin(φ)|where the current waveform is made synchronous with the full waverectified AC line.

The controller (7I10) has to make the moan value of the current waveform(Ipeak*2/π) equal to the average current that is taken by the motor(7I13) and control electronics, from the capacitor C_(M) (7I9).

The current state of the art uses an analog multiplier to multiply theattenuated frill wave AC rectified waveform by the set current for thecontrol loop. (see references 20 and 21).

A proposed method is to sense the time of zero crossing of the AC signaland calculate the phase (φ) of the sinusoidal signal by relating thetime at which a new current value must be set with the period of theline. The period of the line can be measured continually calculating thetime it takes for the number of zero crossing events to occur anddividing the measured time by that number (e.g., 16 is a good choice forthat number).I(t)=I_average_needed*π/2*|sin(ticks_from_(—)0_crossing/ticks_in_period)|Where:

-   -   I_average_needed is the average current required through the DC        motor. ticks_from_(—)0_crossing is the amount of time from the        zero crossing of the AC main measured in an arbitrary unit        (events of a periodic phenomena or ticks_in_period is the period        of the AC main measured in an arbitrary unit (events of a        periodic phenomena or “ticks”);

In this way there is no need for connecting an analog to digitalconverter input of the controlling microprocessor to the AC main.

Table 1 contains a pseudo-code, following the syntax of the C language,showing the algorithm. TABLE 1 // Power Factor Correction Algorithmwithout A/D converter Initialization( ) { P_Ticks = 0 ; // clear ticksaccumulator Ticks = 0 ; // clear ticks accumulator Period = 0 ; // setinitial value of period as not calculated yet Old_Line = 0 ; // setinitial value for algorithm N_cycles = 0 ; // clear 16 cycle counter Settimer interrupt( ) ; // initialise the hardware for a periodic timerinterrupt } // End of initialization Timer_interrupt( ) { // here eachtimer interrupt ; P_Ticks = P_Ticks + 1 ; // count for calculating theperiod Ticks = Tick + 1 ; // count for calculating the phase New_Line =Read_line_state ( ) ; // read state of the AC line − high or low If (New_line <> Old_line ) {   // here if there was a change in state of theAC main   Old_line = New_line ; // save state for detecting nexttransition   If ( New_line = High ) {     // here if a new AC line cycle    Ticks = 0 ; // clear the phase accumulator     N_cycles = N_cycles +1 ; // count the line cycles     If ( N_cycles > 15 ) {     // hereafter 16 line cycles     // a new period average can be calculated    N_cycles = 0 ; // clear cycle counter     Period = P_Ticks / 16 ; //calculate the period average of last 16 cycles     P_Ticks = 0 ; //clear accumulator of next calculation           }         } If (Period > 0 ) { // here to modulate the current      I_out = I * π / 2 *| sin ( Ticks / Period ) | ;     }   else     { // here the first 16cycles, no PFC      I_out = I ;     } // Note: “I” is calculatedelsewhere in a control algorithm suitable for the application } // endof the timer interruptApplication_(—)7I_(—)2. Bilevel CPAP Device

In a bilevel device, the motor needs to take considerable current whileaccelerating to raise the air pressure delivered to the patient. After afew seconds, the motor is braked for decelerating to drop the airpressure. While the current through the motor varies considerably, theaverage current through the inductance needs only to be equal to theaverage current drawn by the motor and the control system (normally thecurrent drawn by the control system is negligible in comparison with thecurrent drawn by the motor).

This application can also make use of the power factor correction methodexplained above.

Application_(—)7K_Power Factor Corrected Brushless DC Motor Controller

The system in FIG. 7K makes use of the topology of FIG. 7J and thetechnology disclosed in FIG. 7I. The DC voltage source (7J01) in FIG. 7Jis equivalent to the output of the full wave rectifier (7K01) in FIG.7K. The capacitance of capacitor Ci (7K02) is of small value so thevoltage in the node X tracks the full wave rectified AC mains waveform.

The functionality of the switch (7K07) is similar to (7J06) and also tothe switch (7I3) in FIG. 7I, however the switch (7K07), like the switch(7J06) is a low-side drive switch, similar to the switch (7A08). Thecurrent sensor (7K08) has a similar functionality to the sensors (7I11),(7J07) and (7A09).

The inductive element (7K06) has similar functionality to (7J05). Thecapacitor (7K05) has a similar functionality to the capacitor (7I9) and(7J02).

The feedback circuit block (7K09) can be implemented by using a pulsewith modulated output from a microcontroller and linking it through anopto-coupler with the set point input of the power factor correctioncontrol (7K03). A practical example of this type of link can be found inFIG. 9H.

The function of the controller (7K03) is similar to the function of thecontroller (7I10) combined with (7I8), as described for FIG. 7I.

The controller (7K03) shapes the current through the inductive element(7K06) so the system has a power factor close to unity.

Any of the current state of the art techniques or algorithms for powerfactor correction can be used in the controller (7K03) (see ref 10 page222 or ref. 25 chapter 1):

-   -   Fix-on time, Discontinuous Current Control (DCC), with or        without fixed output voltage (for the boost topology is called a        “boost follower”)    -   Critical Conduction Mode (CRM) also known as Transitional Mode        Controllers (ref. 25, page 8)    -   Continuous Conduction Mode (CCM) Control (see ref. 20, 21 and        25)

In continuous current control, the control loop can use as feedbackseveral alternatives, for instance, average current control or peakcurrent control.

From a practical implementation point of view, most if not all thecurrent integrated circuits controllers designed for boost converterbased power factor correction circuits can be used with littlemodification for the block (7K03).

If the power factor correction controller is based on a microcomputerwith analog to digital converter the technique disclosed in thedescription of the controllers of FIGS. 7I and 9 can be used.Alternatively, if the factor correction controller is based on amicrocomputer without analog to digital converter the algorithmdisclosed in TABLE 1 can be used.

If a classical controller with fix oscillation frequency, set byexternal components is used along with a microcomputer. The techniqueshown in FIGS. 9H, 9I and 17F, in combination with the algorithm ofTABLE 3, can be used to change randomly the frequency of oscillation ofthe controller, improving on the EMI characteristics of the system.

The block (7K10) is a brushless DC motor controller. (7K12) is thebrushless DC motor. A current sensor (7K11) for the current through themotor (7K12) is shown in FIG. 7K. However, a similar system would workif no direct measurement is made of the current though the motor (7K12).In this case, the current would be assumed by design and adjusted by themain controller (7K14) in response to another measured physical quantityobtained through the sensors (7K15). If there is no need for fastreaction from the brushless DC motor (7K12), the system can be furthersimplified by removing the pulse width modulation from the controller(7K10). That is a slow reacting system still works if no pulse widthmodulation is applied to the windings L1 to L3 of the motor (7K12). Inthis last case, the current through the motor is controlled bycontrolling the average current through the inductance (7K06). Thecurrent through (7K06) tracks the full wave rectified pulsating voltageat the node X. The inertia of the motor plus its load filters thecurrent pulses. This last case is ideal for cooling or ventilation fans,while the full control scenario (usage of sensor (7K11) and pulse widthmodulation of the winding in (7K12)) is ideal for bi-level flowgenerators for the medical industry.

Capacitor (7K13) is included to show that since the controllers (7K14)and (7K10) and the motor (7K12) itself are not connected directly toground, a plurality of de-coupling capacitors with good high frequencyresponse, connected to a ground plane, may be necessary to improve theability of the set-up to pass the electromagnetic compatibilityregulations.

FIGS. 8A to 8C show the relationship between the circuit and thetechnology used in switched mode power supplies (SMPS), as applied toflow generators. To facilitate the comparison, in the three drawings ofFIG. 8, blocks with the same number have a similar function.

FIG. 8A shows a flow generator with a switched mode power supply (SMPS)controlled in voltage mode. The controller 8A1, in the SMPS, usesfeedback from the output voltage of the SMPS (Vout) to control a pulsewidth modulator (PWM). The PWM (85) operates a plurality of switches(82) that interact with a magnetic system (83). The controller (88)controls the current through the BLDCM (86) by applying pulse widthmodulation (8A2) to the electronic commutator (89), in response tofeedback obtained from a physical variable (87).

In FIG. 8B, there is a similar system with the exception of the SMPS.The only change is in the SMPS itself. In fact, the systems in FIGS. 8Aand 8B could interchange the SMPS's provided that both SMPS's meet thesame minimum requirements.

In the current mode SMPS, the feedback from the output voltage is usedby a controller (8B1), but the output of the controller yields the setcurrent (or desired current) of a current mode controller (8B5). Thiscurrent mode controller, may use some common circuitry with someembodiments of the circuit, for instance FIGS. 7E, 7P and 7H. Currentmode control is a superior choice and its advantages for SMPS are wellknown.

The circuit applied to a CPAP flow generator is shown in FIG. 8C. Thereare several blocks in common, but unless a more flexible system like theone already disclosed in FIG. 7I is implemented, the followingdifferences can be seen: D8_1 The voltage over the motor (VM) does notneed to be a controlled variable. The voltage is the consequence of thespeed of rotation. D8_2 The PWM in the electronics commutator (8B2 and8A2) is not required. D8_3 The control system of the power supply itselfis fully integrated into the main controller (88).

Once the flow generator is in operation, a voltage can be measuredacross the motor. Since the control electronics need only a smallfraction (say, as a example, around 5%) of the power needed to operatethe motor, the voltage measured across the motor, as the result of itscontrol operation, can be used to extract the power necessary forcontrol electronics. This is the concept disclosed in FIG. 4 anddetailed further in FIG. 10.

To power the control electronics (44 in FIG. 4 and 10A07 in FIG. 10A)while there is no voltage produced over the motor an “auxiliary powersupply” (42 in FIG. 4 and 10A09 in FIG. 10A) is used. The idea is to usethe “auxiliary power supply” to only provide power during the initialphase of equipment operation after power is applied. Once a suitablevoltage on the motor is established, an electronic switch (43 in FIG. 4and 10A08 in FIG. 10A) is used to unload the “auxiliary power supply”.Alternatively, the “auxiliary power supply” can be shut down by thecontrol electronics itself (10A07) or automatically (see circuit of FIG.10B). If the power drawn by the “auxiliary power supply” is small, andit is acceptable for it to remain operating, a simple current summingjunction can replace the electronic switch (10A08). The advantage ofthis scheme is that the “auxiliary power supply” can be made simply andinexpensively, since the auxiliary power supply operates or drawssignificant power only for a short period of time. As the auxiliarypower supply draws power during a short period of time, the heatgenerated inside the case is small.

There is another form of operation for the system comprised by theelectronic switch and the auxiliary power supply. This system is shownin FIG. 10D.

In FIG. 10D, the auxiliary power supply has a capacitor (10D03) that ischarged with current from the AC main through a resistor (10D01) and arectifier (10D02). The switch (10D05) is normally open. A voltagecomparator with hysteresis, (10D10) senses the voltage in the capacitor.The comparator (10D10) closes the switch (10D05) when the voltage in thecapacitor is larger than a certain value. When the switch is closed, theelectronics (10D12) is powered. Once the electronics are powered, thevoltage in the capacitor (10D03) decreases due to the power consumptionof the electronics. The electronics (10D12) have to activate the motordriver to extract power from the AC main and generate enough voltageover the motor, so a voltage adapter (10D13) can provide power to theelectronics (10D12) via the diode 10D04. The voltage adapter (10D13) istypically a simple linear voltage regulator but can be as complex as aDC/DC converter, depending on the fluctuations and the value of theavailable voltage.

The current state of the art would also use a capacitor like (10D03),charged with a similar system, however, rather than a switch like(10D05), the electronics would be made to operate in a very low powermode. The advantages of this aspect of the circuit are: SW_A1. There isno need for low power electronics in the circuit following the switch.SW_A2. The switch (10D05) and the comparison system (10D06) can beimplemented in a simple way, with commonly available discrete componentsat a lower cost and drawing a lot less power than a system using lowpower electronics. Low power electronics need to draw power to biasinternal circuitry. The differences can be micro-amps, for the low powerelectronics (due to the needs of current biasing circuits), incomparison with nano-amps for the switch (the leakage current of asemiconductor). It has to be noted, that in the current state of the artof low power micro-controllers, the lower power is drawn, only afternormal power is drawn, to initialise the low power mode. Such devicesonly work with the switch option. The resistor (10D01) charging thecapacitor (10D03) is typically never disconnected. The less power thesystem can draw from the capacitor, before operation, the larger theresistor value and the lower the power that is continually dissipated in(10D05). With the possible practical implementations of (10D05) and(10D06), the main problem is the leakage current of the capacitor(10D03) itself. SW_A3. The currently available low power electronicdevices also work from lower voltages than the higher power consumptionoptions. A practical switch can be implemented without that restrictionand switch when a higher voltage is present in the capacitor (10D03).That higher voltage in the capacitor allows the electronics to operatelonger before the other source of power is available.Advantages for an Air-Flow Generation:

The practical advantages of the embodiments of the invention for an airflow generator for CPAP arm:

-   -   (1) Allow a simpler, less expensive power supply as the        requirements of the brush-loss DC motor (BLDCM) and its        electronic commutator are far less demanding than those of the        controlling electronics. This is particularly useful in bi-level        devices as the sudden acceleration of the motor requires extra        power, and power supplies tend to be bigger, more complex and        expensive.    -   (2) Allow the controlling elements of the power supply to be        fully integrated into the controlling electronics (typically        microprocessor based), hence lowering the component count, that        in turn leads to more reliability and lower cost.    -   (3) Allow greater flexibility, since the controlling processor        can be made to control not only the current though the BLDCM,        but also the voltage over the BLDCM that can be made to change        continually as commanded by the control electronics. If as it is        commonly used to control the voltages over the windings, the        electronic commutation is pulse width modulated, it allows a        flexible control scheme in which simultaneously the voltage        available to the pulse with modulating circuit can be changed in        addition to the pulse width itself.    -   (4) If the flexibility discussed above is not necessary, then        the electronic commutation of the BLDCM does not need to include        pulse width modulation. The resulting system is simpler, has        less components and is more efficient, since there is no power        dissipation in the switches (typically MOSFET's) of the        electronic commutator of the BLCDM when performing pulse width        modulation.        Different Implementations        I1. A Complete Flow Generator without Auxiliary Power Supply

FIG. 9 shows an example of an air flow generator for CPAP treatment thatuses the invention disclosed above. A conventional low power AC mainfrequency transformer with two secondary windings is used (9B2 in FIG.9B) to power the control electronics, while the BLDCM (9D2, in FIG. 9D)is driven from the rectified AC main (node VCCLINE in FIG. 9A),following the circuit disclosed (see FIG. 9C) with the topology shown inFIG. 7A. The BLDCM (9D2, in FIG. 9D) Hall effect sensors and itselectronic commutator are powered from the voltage generated fromrunning the motor itself.

FIG. 9A shows the circuit from the AC main IBC power connector (9A6) tothe node VCCLINE. VCCLINE, the output of the circuit, is the full waverectified AC main. This is the DC voltage source (7A01) of FIG. 7A.

Two fuses (9A8) and (9A7) are included for protection. The switch (9A9)is used to turn the equipment on or off. The switched AC main voltage isfound between SA and SN. A metal-oxide varistor (MOV), (9A10) protectsthe device against AC main voltage transients.

A filter (9A1) is used for EMI compliance (reference 8). Therequirements on this filter should be reduced, as the frequency of themain source of EMI is modulated using spread spectrum techniques. Arectifier bridge (9A2) connects as a full wave rectifier, with acapacitive filter provided by (9A4). The capacitor (9A3) improves thehigh frequency impedance of the filter. A NTC (9A5) protects therectifier bridge (9A2) against in-rush current when the equipment isturned on.

FIG. 9B shows the low power transformer (9B2) that provides power to thecontrol electronics. The nodes FMA and FMN in FIG. 9B, are connected tothe input of the rectifier bridge (9A2) after the filter (9A1) and theNTC (9A5). A snubber network 9B1 protects the transformer. One of thesecondary windings is rectified by the bridge (9B3), the output isfiltered by (9B6) and regulated by linear regulators (9B4) and (9B5).This circuit provides +15 Volt to power to the analog sensors, and +5Volt to power the microcontroller. The sensors and the microcontrollerare in FIG. 9J.

FIG. 9C shows the other main components of the circuit. The coil (9C3)is the inductance L_(M), shown as (7A05) in FIG. 7A. The capacitor(7A03) of FIG. 7A is implemented with the capacitors (9C2) and (9C1).Capacitor (9C1) improves the high frequency behaviour of the capacitor(9C2). Diode (9C4) corresponds to diode (7A04) of FIG. 7A. The motor(9C5) corresponds to the motor (7A02) of FIG. 7A.

Electrically, the BLDCM and the electronic commutator (also known asBLDCM as “controller”) is a two terminal network with terminals labelledMOTOR_TOP and MOTOR_BT. The current flowing from MOTOR_TOP to MOTOR_BTis the variable used by the speed control loop, to control the speed ofthe BLCDM and hence the air pressure that the flow generator produces.

The switch of FIG. 7A is implemented by the power MOSFET (9H4) in FIG.9H. The current sensor (7A09) in FIG. 7A is implemented with theresistor (9H5) in FIG. 9H. (7A06) and (7A07) are implemented inside theUC2845 integrated circuit (9H3) in FIG. 9H. The data sheet of the devicemanufactured by Texas Instruments (it used to be a device manufacturedby Unitrode) can be consulted for details. The drain of (9H4) is thenode SW in FIG. 9H that connects to the same node in FIG. 9C. The“desired current” terminal in FIG. 7A is the command information thatthe microprocessor in FIG. 9J sends to the UC2845 in FIG. 9H.

FIG. 9D shows the interface with the BLDCM (9D2). A low voltage droplinear voltage regulator (9D1) gets +5 volts from the terminalMOTOR_TOP. The node is called (+5VSENSORS) in FIG. 9D. A diode (9D27),and a filter consisting of resistor (9D29) and capacitors (9D28) is usedwith the voltage regulator.

The motor sensors (terminals S1M to S3M of 9D2) interface the mainmicrocontroller through optocouplers (9D8, 9D9 and 9D10). The output ofthe optocouplers are the signals S1, S2 and S3, that connect to themicroprocessor (9J2) in FIG. 9J.

Part of FIG. 9D and FIGS. 9B and 9F show an implementation of anelectronic commutator for a typical BLDCM used in airflow generators forCPAP. The circuit shown is particularly flexible in terms of maximum andminimum voltage that the commutator would work with. This is achieved bythe use of a current source to drive the gate of the P-channel MOSFET's.

In FIG. 9D, resistor (9D25), Zener diode (9D26), resistor (9D24) andtransistor (9D23) form a current source. When the optocoupler (9D5) isnot conducting, the current in the collector of (9D25) makes Zener diode(9D11) to conduct, turning the P-channel MOSFET, (9D3) on. Resistor(9D12) is included for protection.

The N-channel MOSFET (9D4) is of logic level type and is turn-on whenoptocoupler (9D6) conducts. A simple resistor (9D13) is shown in thegate of (9D4), however, extra circuitry to accelerate the removal ofcharge from the gate capacitance may be necessary if the pulse widthmodulation of the electronic commutator is used. The point connectingthe drain terminal of both power MOSFET's is connected to L1 terminal ofthe BLDCM, the signal name is L1C.

Negative voltage with respect to MOTOR_BT is used for the currentsource. The voltage is labelled MOTOR_BT_V and is generated from+5VSENSORS by the circuit shown in FIG. 9G. FIG. 9G show a positive tonegative voltage converter using a cascade of two industry standardcharge pump integrated circuits ICL7660 (9G1 and 9G2 in FIG. 9H).

In FIGS. 9F and 9E, the current source, the N and P channel powerMOSFET's and the two optocouplers have a function similar to the circuitin FIG. 9B that drives the L1 winding of the motor. The L2 winding ofthe motor is driven by circuitry in FIG. 9E, while the L3 winding of themotor is driven by circuitry in FIG. 9F.

FIG. 9H shows a practical circuit that controls the switch (7A08) inFIG. 7A. An integrated circuit normally found in off-the-main switchedmode power supplies, the Unitrodo's UC2845 (9 h 3) is controlling apower MOSFET (9H4). The current sensor (7A09) of FIG. 7A is the resistor(9H5). RC network (9 h 7) and (9 h 8) is a low pass filter that isnecessary to eliminate switching noise from the voltage resulting firmsensing the current.

The voltage of one of the windings of the transformer (9B2) is rectifiedby (9 h 1), filtered by (9 h 12) and regulated to 12 volts by (9 h 2).The 12 volts are used to power the current mode controller chip (9 h 3).

Transistor (9 h 12) saturates due to the current through resistor (9 h9), connected to the +12 volt supply. A low voltage in pin 1 of (9 h 3)inhibits the operation of the chip and keeps the power MOSFET (9 h 4)off. To start the operation of the chip, or to make it work in bursts,to achieve very low currents through the motor, the main microprocessoroperates the optocoupler (9 h 10) through resistor (9 h 11). A low levelin the terminal labelled (shdwn) enables the operation of the currentmode controller. In this way, the initial current can be set, before thechip can operate.

The current is set through pulse width modulation applied to resistor (9h 20), connected to the optocoupler (9 h 19). The waveform in thecollector of (9 h 19) is filtered by resistor (9 h 16) and capacitor (9h 17) and is buffered by operational amplifier (9 h 15) to give avoltage that is used by the error amplifier of the UC2845 (9 h 3). Theerror amplifier is programmed with a gain of (−1) by resistors (9 h 14)and (9 h 13). Capacitor (9 h 21) limits the bandwidth of the amplifier.The voltage in pin 1 of the UC2845 is reduced internally by subtractingtwo diode drops and divided by three before being compared by thevoltage sensed by the pin labelled (CS). When the voltage in (CS)exceeds the voltage in pin 1, as processed above, the switch (9 h 4) isturned off. The switch (9 h 4), is turned on by the oscillator insidethe UC2845.

The optocouplers (9 h 10) and (9 h 19) are controlled by themicrocontroller (9J2) that controls the main function of the airflowgenerator. By changing the parameters of the pulse width modulation inthe optocoupler (9 h 19), the main microcontroller changes the maximumcurrent through the sensing resistor (9 h 5), hence changing the averagecurrent through the BLDCM. The other optocoupler, (9 h 10) can be usedto set the current to “0” for several cycles, hence allowing for smallaverage current to flow through the BLCDM. This optocoupler can also beused in case of error or when the equipment is in stand-by.

The timing for the oscillator is provided by a RC network connected tothe pin labelled (RC) in (9 h 3). Normally a simple RC network isconnected there.

FIG. 9I shows a circuit to change the frequency of operation of theUC2845, under the control of the microprocessor (9J2). This is done touse spread spectrum techniques to facilitate EMI compliance (reference8). Pulse width modulation is applied through the optocoupler (9I2). Thewaveform in the emitter of the output transistor of (9I2), is filteredby (9I4) and (9I6). (9I3) and (9I5) form a resistive divider that limitsthe maximum voltage over the filter. The output is referred to +Vref,the 5 volts reference in the UC2845. Operational amplifier (9I7), andtransistor (9I12) are connected as a current source that is proportionalto the voltage between +Vref and the output of the filter (voltage overthe capacitor (9I6)). This current charges the capacitor (9I13) that ispart of the oscillator of UC2845. Resistor (9I10) provides a minimumcurrent for charging the capacitor, giving a minimum frequency to theoscillator.

Another way of changing the frequency of the oscillator is to replacethe resistor in the RC oscillator of the UC2845 with a digitallycontrolled potentiometer.

FIG. 9-J shows the main microcontroller of the flow generator. Themicrocontroller (9J2) operates the electronic commutator of the BLDCM(9D2) and the user interface (9J3) and reads the information fromsensors (9J4). The circuit block (9J4) represents the pressure andairflow sensors typically found in the top end CPAP flow generators.(9J2) also interfaces with the current mode controller UC2845 of FIG.9H. This microcontroller could also communicate with other clinicalequipment (typically through serial lines) as it is well known practice.

The microcontroller (9J2) sets the current through the BLDCM by usingthe pulse width modulator PWM2 and an output line (signal name SETC andshdwn) connecting to the circuit of the UC2845. (9J2) also changes thefrequency of operation of the UC2845 by using another pulse widthmodulator (PWM3). The microcontroller (9J2) in FIG. 9J has to generatethe needed dead time. That is, the time necessary to prevent bothMOSFET's, the N-channel mid the P-channel to conduct at the same time,effectively short-circuiting MOTOR_TOP with MOTOR_BT.

A voltage to frequency converter, appearing as a block (9J6) in FIG. 9J,connects to microcontroller (9J2) through optocoupler (9J7). This isused to measure the voltage between MOTOR_TOP and MOTOR_BT.

FIG. 9K shows a circuit for the voltage to frequency converter (9J6).The voltage in MOTOR_TOP is filtered and regulated by a low voltage drop5V linear regulator (9K1). Operational amplifiers (9K16) and (9K22)along with transistors (9K21) and (9K6), form a controlled currentmirror. The current that charges the timing capacitor (9K8), depends onthe voltage across resistor (9K15), that in turn, depends on the inputvoltage, the voltage between MOTOR_TOP and MOTOR_BT. The capacitor (9 k8) is connected to a CMOS 555 timer IC, (9K9). The capacitor (9 k 8) isdischarged by resistor (9K7), which is operated by the timer IC.Discharge time is constant and the charge time depends on the inputvoltage. Optocoupler (9K11) in FIG. 9K corresponds to Optocoupler (9J7)in FIG. 93.

A crowbar circuit (11A06), as shown in FIG. 11A, may be added, connectedbetween MOTOR_TOP and MOTOR_BT as a protection device. The crowbar is awell known circuit in which a SCR is triggered when the voltage acrossthe terminals of the circuit exceeds a given value, working as aprotection against malfunctions. Usually when a crowbar is triggered, afuse (11A04 in FIG. 11A) is blown to prevent further damage.

An alternative way of operating the electronic commutator of the BLDCMis to use a “small”, 20-pin microcontroller (for instance, the AT90S2313from Atmel Semiconductor) to perform the logic and interface with themain controller via an optically coupled serial NRZ fall duplex line.This solution releases the main microprocessor from the task ofoperating the electronic commutator. The 20-pin microcontroller canreport data to the main controller, like speed of the BLDCM, as it wouldbe connected to the motor sensors. Commands to brake the motor oroperate coasting can be given by the main microcontroller to the 20-pinmicrocontroller through the serial communication line. Also withadditional interface circuitry, the “small” 20-pin microcontroller canreport the voltage over the motor or can operate its own pulse widthmodulator or current control circuitry, to have a system as it wasdisclosed in FIG. 7I. The second “small” microcontroller can also beused as a complex watch dog timer. Both microcontrollers can reset eachother in case of detection of a fatal error condition.

FIG. 9L shows a low voltage drop linear regulator suitable to be used in(9D1), (9K1), and (11A17). The circuit is designed to be used with 24volt motors, since currently available integrated circuits like, forinstance, the LM2940, can work continually only with 20 VCC.

In the circuit of FIG. 9L, an operational amplifier (9L21) works as theerror amplifier of a linear voltage regulator.

Resistors (9L26) and (9L27) form the feedback network, and itsrelationship determines the output voltage. (9L28) is a voltagereference that is powered from the output of the circuit itself throughresistor (9L24). A PNP topology is chosen for the pass transistor of thelinear regulator (9L5) to obtain low voltage drop. Transistor (9L7) andresistor (9L6) provide current limiting. The output current circulatesthrough resistor (9L6) and when the voltage drop reach a value thatmakes the base-emitter junction of (9L7) to conduct, the current throughthe collector of (9L7) diverts current from the base of (9L5), limitingthe output current.

Resistors (9L9) and (9L8) bias transistor (9L5) from the drain terminalof a logic level MOSFET (9L13). The gate terminal of the logic levelMOSFET (9L13) is connected to a voltage source implemented with 3.3 voltZener diode (9L11), through resistor (9L12). Resistor (9L10) providesbias current to the 3.3 volt Zener diode (9L11). The collector currentof transistor (9L14) can cause a voltage drop in resistor (9L12) andturn off the logic level MOSFET (9L13).

On powering up the circuit, when the output voltage is lower than theminimum voltage needed for the operation of the operational amplifier(9L21) and transistor (9L17), the logic level MOSFET (9L13), issaturated as its gate has the full 3.3 volts of the Zener diode (9L11).In this condition, maximum current circulates through the resistor(9L8), from the base of transistor (9L5). The current charges outputcapacitors (9L22) and (9L23). This arrangement guarantees that thecircuit powers up. When the operational amplifiers senses that thevoltage at the output, us reduced by the resistive divider (9L26) and(9L27) is larger than the voltage reference (9L28), the voltage on itsoutput is raised.

The output of the operational amplifier (9L21) is connected to aresistive divider formed by (9L18), (9L17) and (9L16). The resistivedivider formed by (9L18), (9L17) and (9L16) biases the base oftransistor (9L17). Because of the emitter resistance (9L15), transistor(9L17) works as a current source. The higher the voltage differencebetween the sampled circuit output voltage in the non inverting (+)input of the operational amplifier (9L21), and the reference voltage(9L28), connected to the inverting input of (9L21), the higher thevoltage at the base of (9L14) and the higher the current through itscollector.

The higher the current through collector of (9L14), the lower thevoltage in the gate of logic level MOSFET (9L13), and the lower thecurrent through the base of (9L5). The lower the current through thebase of (9L5), the lower the current through its collector, and thelower the output voltage of the circuit is, compensating the originaleffect. Capacitor (9L19) and resistor (9L20) compensate the feedbackloop, by shaping its frequency response. Resistor (9L1) and capacitors(9L4) and (9L3) form a low pass filter for the noise and voltagevariations of the input voltage V_(M). Schottky diode (9L2) is needed ifthe voltage available from V_(M) is discontinuous.

I2. Auxiliary Power Supply DC-DC Converter and Switch

FIG. 10B shows an example of the “auxiliary power supply” referred to inFIG. 10A and FIG. 4. This power supply typically dissipates significantpower during the first seconds of operation of the unit. It consists ofthe shunt regulator (Z2), formed by a programmable Zener diode TL431,that is followed by a power MOSFET (MF1) connected to the voltagegenerated by the TL431 via the resistor (R3). The combination of Zenerdiode (Z1), transistor (T1) and the resistors (R6) and (R7), makes thevoltage at the output to stabilise around 12 volts.

If the voltage at the output (point labelled APS) grows, the basecurrent of transistor (T1) also grows, causing the collector current oftransistor (T1) to grow. This, in turn, causes the voltage in the gateof MOSFET (MF1) to decrease, making the output voltage to decrease. Inthis way, the system is stable. The electronic switch shown in FIG. 4,is made with diodes (D1) and (D2) for sake of clarity.

When the linear regulator 7815 produces 15 volts, diode (D2) conductsand diode (D1) is reverse biased. However, in a practical circuit, diode(D1) can be replaced by a short circuit, since a voltage larger thanaround 13 volts in the output (node labelled APS) makes the transistor(T1) conduct in such a way as to cut off MOSFET (MF1). This leaves onlya small current flowing every half cycle of the AC main through theprogrammable Zener integrated circuit TL431 (Z2). The voltage togenerate the 15 volts of the 7815 regulator is generated from thevoltage over the motor by a suitable DC-DC converter.

Alternatively, the inductance L_(M) ((64) in FIG. 6) connected to theBLDCM can be replaced by a transformer, with one of its secondarywindings generating the voltage for the 7815.

I3. DC-DC Converter

FIG. 10C shows an implementation of the DC-DC converter necessary toextract useable voltage out of the motor voltage.

The DC-DC converter is designed to be controlled from the mainmicroprocessor. The design can be used to reduce or boost the voltagecollected from MOTOR_TOP, according to the mode of operation. Thecircuit can operate in three modes: DCDC_m1: As a down converterswitched mode power supply. The voltage from MOTOR_TOP is filtered byresistor (10C01) and capacitors (10C02) and (10C03). The transistor(10C04), diode (10C06) coil (10C05) and capacitors (10C08) and (10C09)can operate as a down converter switched mode power supply, when motorvoltage is high. In this mode of operation the MOSFET (10C12) is off andtransistor (10C04) is controlled by a pulse width modulator output fromthe microcontroller (PWM_PS). The microprocessor uses the voltagesampled by (10C11) (node labelled MPADPS) to close the loop. DCDC_m2: Asa boost converter switched mode power supply. The voltage from MOTOR_TOPis filtered by resistor (10C01) and capacitors (10C02) and (10C03).Transistor (10C04) is saturated all the time. The MOSFET (10C12), diode(10C07) coil (10C05) and capacitors (10C08) and (10C09) operate as aboost converter (also called a “ringing choke”) switched mode powersupply, when motor voltage is low. In this mode of operation, the MOSFET(10C12) is controlled by a pulse width modulator output from themicrocontroller (PWM_PS). The microprocessor uses the voltage sampled by(10C11) (node labelled MPADPS) to close the loop. DCDC_m3: As a linearpower supply. The voltage from MOTOR_TOP is filtered by resistor (10C01)and capacitors (10C02) and (10C03). Transistor (10C04) is saturated allthe time. The MOSFET (10C12) is off all the time. The coil (10C05) andcapacitors (10C08) and (10C09) act as further filtering. Diode (10C07)conducts all the time.

The microprocessor selects the mode of operation based on the voltageover the capacitor (10C08), by operating the output line PS_MODE. Theline PS_MODE, changes the way analog switches (10C14) and (10C13)connects

I4. Start-Up Switch

FIG. 10 shows an implementation of the switch (10D05) of FIG. 10D andthe circuit that operates the switch (10D06) in FIG. 10D. Whentransistor (10E13) is off, and the voltage at the input is lower thanthe Zener voltage of diode (10E10) plus the minimum voltage drop forconduction in the base of transistor (10B13), the transistor (10E03) isoff. The current taken from the input voltage is the leakage current ofall the transistors connected. When the voltage at the input is highenough for the base of (10E13) to conduct with current through the Zenerdiode (10E10), transistor (10E03) is turned on, as collector of (10E13)draws current and is pulled down. Also, transistor (10E06) is turned on.

The collector current of (10E06) goes through the base of (10E13)establishing a positive feedback loop, that keeps (10B13) conductingeven when the input voltage Vi becomes lower than the minimum voltagerequired to start the operation of the switch.

Transistor (10E04) and resistor (10E02) limit the current through thetransistor (10E03). In this way, large capacitors can be connected to Viand Vsw, without damaging the transistor (10E03).

I5. Flow Generator with Auxiliary Power Supply

FIG. 11A shows an air flow generator using the concept of auxiliarypower supply.

The topology is similar to that of FIG. 6 with the switch implemented byMOSFET (11A05), the DC power source the rectified AC main as in FIG. 9A(note node VCCLINE). The inductance is coil (11A08) and the freewheelingdiode is (11A07). The capacitor in parallel with the BLDCM is (11A11).The current is sensed by resistor (11A14). The BLDCM is (11A12).

All the control circuit runs from +5 volt from a low voltage dropregulator (11A17). The auxiliary power supply (11A01) can be implementedwith the circuit of FIG. 10B, modified to give around 6 volt at theoutput rather than 12 volt. This is done by changing Zener diode (Z1) inFIG. 10B to a 5.6 Volt Zener diode rather than a 11 Volt one.

An alternative to the auxiliary power supply (11A01) is the arrangementof FIG. 10D. With the circuit of FIG. 10E replacing the switch (10D05)and the circuit block (10D06), the low voltage drop SV regulator (11A17)and all the circuits running from 5 Volt take the place of (10D12), andthe voltage adapter (10D13) is resistor (11A02) and capacitor (11A03).

The microcontroller (11A16) has inputs and outputs that can interfacewith a brush-less DC motor and electronic commutator similar to the oneshown in FIG. 9. Consequently, the control lines out of (11A16) havebeen labelled consistently with the output lines out of (9J2). Thevoltage in the node MOTOR_TOP is monitored using an analog input to themicrocontroller (A/D converter) through network (11A15). The MOSFET(11A05) is driven by a pulse transformer (11A09). The electroniccommutator of the BLDCM can be pulse width modulated to enable a moreflexible operation. The effect is achieved by applying one of the PWMoutputs of the microcontroller to the gate (11A18).

FIG. 11B shows a current mode controller designed to work % with thecircuit of FIG. 11A. One of the pulse width modulator output lines ofthe microcontroller clocks the “D” type flip-flop (11B05), setting theoutput to one. When the current reaches the level set by another pulsewidth modulator, through filter formed by (11B12) and (11B13), thecomparator (11B06) resets the flip-flop. The output of the flip-flop isconnected to buffers that drive the transformer (11A09). Gate (11B04)and inverter (11B01) provide fail-safe operation. If the line labelledCDFF from the microcontroller is low, the output buffers are disabledand the flip-flop cannot be set to one.

The current is measured also by the microcontroller by reading the A/Dinput connected to the operational amplifier (11B1), which is amplifyingthe voltage sensed by the resistor (11A14). A crowbar circuit is shownin FIG. 11A (11A06) it is connected between MOTOR_TOP and MOTOR_BT as aprotection device. The crowbar is a well known circuit in which a SCR istriggered when the voltage across the terminals of the circuit exceeds agiven value, working as a protection against malfunctions. When thecrowbar is triggered, the fuse (11A04) is blown to prevent furtherdamage.

I6. Multiple Cooling Fan Controller

FIG. 12 shows another application of the embodiments of the invention,as disclosed in FIG. 7H. A transformer (1203) connected as in a flybackdc-c converter is used to provide the inductance and the freewheelingdiode. In the circuit of FIG. 12, up to four 12 volt cooling fans can beconnected in series. This limit of four, is to limit the maximum voltageat the output of the circuit. Each of the cooling fans is connectedthrough a connector and can be replaced by a link when not connected.

A current mode controller (1210) is used to control the MOSFET (1208)while sensing the current with resistor (1205).

The transformer has an additional winding to provide power to thecurrent mode controller through voltage regulator (1202). The circuitblock (1201) provides the power for starting the current modecontroller.

In the secondary side of the transformer, a microcontroller (1212) ispowered from voltage regulator (1205). (1205) takes the power from thevoltage resulting from the cooling fans when the set output current (Io)is circulating).

Resistor (1218) is used to sense the output current. The microcontroller(1212) can sense the voltages V0 to V3 in the circuit, corresponding tothe voltage across each of the cooling fans, and the top of the stack.Each fan has a capacitor in parallel, as shown in the block (1204).Temperature can be sensed by a sensor (1217) and read by an analog inputof the microcontroller. The microcontroller interacts with the currentmode controller through an isolated interface, as in FIGS. 9H and 9I, tochange the current (Io) and change the frequency of operation of thecurrent mode controller randomly to facilitate EMI compliance.

An isolated mains AC zero crossing circuit may be added to themicrocontroller, to allow for the current (Io) to be changed as a fullwave rectified AC signal synchronous with the AC main, in order toimprove the power factor of the circuit.

(1206) and (1207) are snubbers, which may be necessary in any flybackcircuit, depending of the characteristics of the transformer used.

I7. Cooling Fan

FIG. 13 shows a cooling fan based on a brush-less DC motor running fromthe AC main with the system shown in FIG. 7A. A rectifier bridge (131)rectifies the AC main. The inductance (L_(M)), (135) corresponds withthe inductance (7A05) of FIG. 7A, while the capacitor in parallel withthe motor C_(M) (134) corresponds with (7A03) of FIG. 7A. The anode ofthe freewheeling diode (1350) is connected from the drain of powerMOSFET (137) and the cathode of the freewheeling diode is connected toVCCLINE.

The network consisting of (1314), (1313) and (1315) provide power tostart a UCC3845 based current mode controller. Resistor (1322) issensing the current through the switch (137). In operation, charge is“pumped” through the circuit consisting of capacitor (136), diodes (139)and (138) and Zener diode (1310). The voltage over the Zener diode(1310) is connected through diode (1311), to capacitor (1312).

A microcontroller (1318) powered from a low voltage drop regulator(1323) is used to perform the following tasks: Task_Fig13_1 Measure thevoltage across the fan to check its operation and to have an indicationof its speed. Task_Fig13_2 Set the current through the cooling fan byInterfacing to the (1317). It can be modulated to correct power factorby measuring the voltage VCCLINE and making the current through the coilto be a full wave AC rectified waveform synchronous with the voltage inVCCLINE. If this is done, capacitor (132) is a small capacitor, allowingthe voltage at the output of the rectifier to track the AC line fullwave rectified voltage. Task_Fig13_3 Change the frequency of operationof (1317) following spread spectrum techniques to facilitate EMIregulatory compliance. (reference 8). Task_Fig13_4 Control the speed ofthe fan. Task_Fig13_5 Interface through optocoupler (1321) to atemperature to frequency converter, to set the speed of the fan inresponse to changes in ambient temperature.

The circuit of (1317) can be taken from FIGS. 9H and 9I, with the powertaken from capacitor (1312), through diode (1316) instead of the 12 voltregulator (9H2) of FIG. 9H.

I8. Bilevel Flow Generator with Power Factor Correction

FIG. 14 shows an application of an embodiment of the invention disclosedin FIG. 7I.

The AC main voltage is passed through a circuit similar to the circuitshown in FIG. 9A. The filtered AC mains voltage is rectified by bridge(141). Capacitor (142) is small enough for the voltage across it totrack the full wave rectified AC main.

Switch (1414) is a transformer driven power MOSFET, similar to theswitch of FIG. 11. Inductance (1415) has a function similar to (715), afreewheeling diode (1420) is connected as (714) in FIG. 7I.

The motor driver (143) use the pulse width modulation of the electroniccommutator to drive motor (144) in constant current mode, using sensor(146).

Controller (1412), control the current through the inductance (1415),that is approximately the current through the sensor (1417).

The current through the motor and the current through the inductance areset by the main controller (149).

Sensors (148) translate physical variables into information that (149)can read.

The system in FIG. 14 can operate in several modes as follows: F14M_1.If the microcontroller measures the voltage in the point connectingC_(M) and L_(M), and use the measurement as feedback in a control loopthat changes the current through the coil (1415) to obtain the desiredvoltage over the motor, then the system operates as the current state ofthe art. F14M_2. If the system operates as described above, but thevoltage is changed to assist the action of the motor driver (143), thesystem is behaving in an innovative way. For instance, the voltage maybe raised immediately before the motor needs acceleration. Or thevoltage may change to make the action of the pulse width modulation moreeffective. F14M_3. The system can work as disclosed in FIG. 7I. Theaverage current that the motor controller (143) uses is allowed to passthrough inductance (1415). Charge is accumulated in C_(M), to be usedwhen peak current is needed. F14M_4. This mode is similar to theprevious mode (3), but the current through inductance (1415), (L_(M)) isshaped as a full wave rectified sinusoidal waveform, synchronous withthe waveform in the rectified AC mains, to improve power factorcorrection. This can be achieved either by multiplying the currentoutput by the value read in the analog to digital converter inputthrough network (1410) or with the method described in with thedescription of FIG. 7I. The network (1411) connects the microcontroller(149) to the filtered AC mains, for zero crossing detection. The resultis shown immediately below (1411) in FIG. 14.

Current sensor (146) does not need to be used, and the pulse widthmodulation of the microcontroller (149) can operate directly over theelectronic commutator of the BLDCM (144), in response to some physicalquantity that is controlled. Still similar results can be achieved.

The feedback loop controlling the average current through the inductance(1414) has to use the voltage over the motor, to calculate the actualcurrent draw by the motor itself. This is done by the calculation:Current_in_or_out_of_(—) CM=CM*Diff_in_voltage/Time_between_samplingandI_motor=I_inductance−Current_in_or_out_of_(—) CM

where: Diff_in_voltage: The change in capacitor voltage since the lastsample. Current_in_or_out_of_CM: The net current through the terminal ofthe capacitor. Time_between_sampling: The time between the samples madeto the voltage over the capacitor. CM: The value of the capacitor inparallel with the motor. I_motor: Average current through the motorduring sample interval. I_inductance: Current through the inductance ascontrolled. It is measured in the sensor (1417).

Current state of the art feedback loops, to control the voltage ofoutput of a power supply, use an error magnitude that would be thedifference between the voltage over the capacitor and the desiredvoltage. If the difference is large enough, the output of the controlloop will saturate.

In this new control scheme, even large voltage differences between thecapacitor instantaneous voltage and its average voltage do notnecessarily cause a similar response; the response depends on thecurrent requirements.

Table 2 lists pseudo-code following the syntax of the C languagedetailing the algorithm.

If the system also corrects for power factor, the sampling interval is 1cycle of the AC main. The sampling interval can be made half the cycleif the extra harmonics introduced by occasionally having differentcurrent in both half cycles of the AC mains is acceptable. TABLE 2 //algorithm to determine the value to set the current through theinductance (LM) // according to the charge stored in the capacitor inparallel with the motor (CM) VC =reading_capacitor_voltage_from_AD_converter() ; if ( VC > VCmax ) { I =0 ; } // check that the voltage is not too high else {   If ( VC = VCmin) { I = IMAX ; } // check that the voltage is not too low    else   {  V_Diff = VC− Old_VC ; // calculate the voltage difference betweensamples   // the two lines bellow are necessary if the average currentthrough the   // motor is unknown   I_cap = C * V_Diff /Time_between_samples ; // calculate the net current // into thecapacitor   I_motor = I_inductance − Icap ; // calculate the currentthough the motor   If ( VC > V_target )     {     // here if there is noneed for accumulating additional charge     I = I_motor ;     }   else    {  // here if the capacitor voltage is less than or equal to thetarget voltage     // calculate the current needed to replenish thecapacitor     I_to_charge_C = ( V_target − VC ) * C /Time_between_samples ;     // add the current needed to replenish to thecurrent needed by the motor     I = I_motor + I_to_charge_C ;     }   }} Old_VC= VC ; // save the current value of the capacitor voltage Return( I ) ; // return the calculated current // end of the function

Once the average current through the inductance is calculated, itremains constant for the sampling time. If the system is power factorcontrolled, the average current through the inductance is used tocalculate the peak of the full wave rectified sinusoidal current that isapplied through the inductance, synchronically with the AC mains as:I(t)=I_average through_inductance*π/2*|sin((2*π*F)*t)|where:

-   -   I(t) is the set current to the current mode controller.    -   I_average through_inductance is the current requirements keeping        the motor running and the capacitor with the target charge.    -   F is the frequency of the AC main.    -   t is the time.

The operating frequency of the current mode controller, is changed usingthe algorithm shown in Table 3. TABLE 3 Algorithm for changing thefrequency randomly to improve EMI compliance // Table of frequency dataFT[0]= Data frequency 0; FT[1]= Data frequency 1; FT[2]= Data frequency2; . . . . . . . . . . . . . . FT[N−1]= Data frequency N−1 ; IntGet_new_frequency(Old_frequency) { new_number: R =generate_random_number_between_N−1_and_0 ( ); If ( Old_frequency = 0 )and ( R=N−1 ) { goto new_number ; } If ( Old_frequency = N−1 ) and ( R=0) { goto new_number ; } If ( R = Old_frequency ) { goto new_number ; }If ( R = Old_frequency − 1 ) or ( R= Old_frequency +1 ) { gotonew_number ; } Return ( R ) ; } // note the hardware is programmed withdata from the table using the number as index.FIG. 16 a Topology for an Active Power Factor Correcting Circuit

FIG. 16 shows a topology for an active power factor correcting circuit.Although related to other switch mode power supply topology's, thecircuit is unique in the fact that the output is not referred to theinput ground, but rather to the input voltage.

The circuit could be described as “a non-isolated flyback switcher withthe output referred to the input”, or “a low side switched, buck-boostconverter with the output referred to the input”. It is also related tothe boost converter. The buck-boost converter is the switch mode powersupply topology from which the circuit of FIG. 7L is derived from.However, unlike the boost converter, the output voltage can be sethigher or lower than the input voltage (the rectified AC mains).

In FIG. 16, a diode bridge (1601) rectifies the AC mains into a fullwave rectified AC waveform. The value of capacitor C_(SMALL) (1602) hasto be chosen so the voltage in the point “A” of the circuit (thepositive output of the diode bridge rectifier (1601)), tracks thevoltage produced by the rectifier. The output of the circuit of FIG. 16is the port from “A” to V_(O), consequently the point “A” has been madethe input common terminal for the main converter (1607) in FIG. 16. Themain converter (1607) is what it is normally call the “load” of thecircuit of FIG. 16.

Capacitor Co (1606) is a storage capacitor. It is in parallel with theoutput. The switch, (1605) in FIG. 16 is typically an N-Channel powerMOSFET. When the switch is closed, the inductive element L_(PFC) (1603)is connected across the input voltage and current flows into it, growinglinearly during the period of time the switch is conducting (T_(ON)). Atthe frequency of commutation normally used in this type of circuits, thefull wave rectified AC mains remains practically constant for the periodof commutation. The controller (1610) in FIG. 16 uses the informationprovided by the current sensor (1608) to decide when to open the switch(1605). When the (1605) opens, the diode (1604) takes the currentcirculating from the inductive element (1603). The current through theinductive element can not change instantaneously and starts flowing intoboth the load and the storage capacitor Co (1606) connected in parallelwith the load. With the switch (1605) open and the diode (1604)conducting, the inductive element (1603) is just in parallel with theseries combination of the diode (1604) itself, in series with theparallel combination of Co (1606) and the load (main converter 1607). Ifthe switch is kept open, current keeps flowing, charging capacitor Co(the portion of the current not taken by the load) until the currentthough the inductive element reaches zero. At this moment the diode(1604) becomes reverse bias by the voltage across the storage capacitorCo.

If the switch closes before the current reaches zero, then the circuitis said to operate in the continuous current mode. The process startsagain in the next cycle. By the way the Power Factor Controller (PFC)(1610) controls the maximum value of the current trough the inductiveelement (1603), and the conduction time of the switch (1605), thefiltered waveform of the train of triangular (or trapezoidal for thecase of continuous current mode) current pulses taken from capacitor(1602) is filtered by the action of the capacitor into a normally goodapproximation of full wave rectified AC mains current waveformsynchronous with the AC mains, that yields close to unity power factorwhen “seen” from the AC mains side. The feedback circuit (1609) isneeded to provide the controller (1610) with the information of thevoltage across the storage capacitor Co (1606).

This topology can be seen as related to the system of FIG. 7J, that hasbeen analysed in FIG. 15.

The analysis is the system in steady state is similar to the analysismade in FIG. 15. The “load” of the system is the block (1607). Any typeof DC-to-DC or DC-to-AC converter could be used in block (1607).Although FIG. 16 shows a converter, it will be obvious for those skilledin the art that any load suitable to be connected to a power supplycould be used in place of block (1607).

Consequently with the results of FIG. 15, it can be said that fromdynamic point of view, the circuit can be designed using the well-knownequations of the buck-boost converter (also known as positive tonegative converter). However, it must be kept in mind that the output isreferred to the input voltage and not to ground, the polarity of theoutput is not negative, and that the switch must be rated as the sum ofthe input voltage plus the output voltage.

The advantages of this topology are:

-   -   Low side switch (unlike the buck-boost or positive to negative        converter and the buck converter topology)    -   No in-rush current problem (unlike the boost converter)    -   High power factor correction (unlike the buck converter        topology)    -   Freedom to choose the output voltage. Normally it will be lower        than the output voltage to be able to simplify the converters        following the circuit. However the output voltage could be made        variable.

In the classical non-power-factor-corrected topology for an off themains switch mode power supply implemented as a buck converter followedby a DC-DC converter operating from the lower voltage created by thebuck converter, it would be an advantage to replace the buck converterby the circuit of FIG. 16 to give power factor correction. However, asit would be clear for a designer skilled in the art, it must be notedthat for optimum performance the DC-DC converter following the circuitof FIG. 16 must be itself controlled with its own feedback loop from theoutput of the power supply.

The advantages of lowering the voltage obtained from the power factorcontroller are:

-   -   P type MOSFET's are rarely manufacture for drain to source        breakdown voltages voltages over 200 Volts. P channel MOSFET's        permit more flexible designs, with simple drivers.    -   MOSFET's with lower drain to source breakdown voltages may be        faster.    -   MOSFET's with lower drain to source breakdown voltages has lower        ON resistance.    -   Logic level MOSFET's are rarely manufacture for drain to source        breakdown voltages over 60 Volts.    -   Working with lower voltages makes design and prototyping easier.    -   The voltages of many electronic circuits are very low. Hence, it        is an advantage to drive a high performance DC-DC converter from        a lower voltage.

For switched mode power supplies (DC to DC converter following thecircuit of FIG. 16) or other devices like fluorescent tubes drivers (DCto AC converter following the circuit of FIG. 16) the main converterfollowing the power factor corrector stage is isolated hence the factthat the output of the circuit of FIG. 16 is not referred to ground is aproblem only from the point of view of noise and for the complexity ofthe feedback circuit. Both problems can be overcome.

The disadvantage of not having the output referred to ground can beeasily overcome by decoupling carefully with capacitors with good highfrequency characteristics to form a ground plane at the frequency of thecommutation of the converter.

The feedback from the output of the controller has to be done by eitheran isolated link or using a differential amplifier. The most inexpensivetechnique would typically be to use an optocoupler. However, all thetechniques currently used for providing feedback though an isolatingbarrier can be used. Alternatively, a differential amplifier can be usedtaking the input from the two output terminals of the circuit of FIG.16. This eliminates the input voltage from the mains. This voltageappears as common mode voltage between the output terminals of thecircuit.

A topology based on a buck-boost converter may use a simple voltagedivider for the feedback circuit, and the output voltage can be lowerthan the input voltage, however, the requirement of a high side drivefor the switch far out-weights the advantage of the simpler feedbackcircuit.

Any of the current state of the art techniques or algorithms for powerfactor correction can be used in the controller (1610) (see ref. 10 page222 or ref 25 chapter 1):

-   -   Fix-on time, Discontinuous Current Control (DCC), with or        without fixed output voltage (for the boost topology is called a        “boost follower”).    -   Critical Conduction Mode (CRM) also known as Transitional Mode        Controllers (ref. 25, page 8).    -   Continuous Conduction Mode (CCM) Control (see ref 20, 21 and        25).

In continuous current control, the control loop can use as feedbackseveral alternatives, for instance, average current control or peakcurrent control.

From a practical implementation point of view, most if not all thecurrent integrated circuits controllers designed forboost-converter-based power-factor-correction circuits can be used withlittle modification for the block (1610) of FIG. 16.

If the power factor correction controller is based on a microcomputerwith analog to digital converter the technique disclosed in thedescription of the controller of FIGS. 7I and 9 can be used.Alternatively, if the factor correction controller is based on amicrocomputer without analog to digital converter the algorithmdisclosed in TABLE 1 can be used.

If a classical controller with fix oscillation frequency, set byexternal components is used along with a microcomputer. The techniqueshown on FIGS. 9H and 9I and FIG. 17F in combination with the algorithmof TABLE 3, can be used to change randomly the frequency of oscillationof the controller, improving on the EMI characteristics of the system.

FIG. 17 Application of Topology for Power Factor Corrected AC to DCConverter of FIG. 16

FIG. 17 shows a practical example of one possible implementation of thetopology shown in FIG. 16. The NCP1650 power factor controllerintegrated circuit, from On Semiconductors (see ref. 26), has beenchosen for the example. The NCP1651 (see ref. 27), from the samemanufacturer, is a more suitable device since it does not require theexternal start-up circuitry and the feedback input is specificallydesigned to work with the optically coupled circuit used in this type ofapplications. However the NCP1650 has been selected instead of theNCP1651 to show that most if not all power factor correction integratedcircuits can be adapted to work with the topology of FIG. 16.

As with all the integrated circuits, its application must comply withthe design parameters defined by the manufacturer. The example of FIG.17 is a variation of the following application note from Onsemiconductor: AND8106/D, 100 Watt, Universal Input, PFC Converter. Itcan be found on page 67 of reference 25. The differences with theoriginal application note are:

-   -   The topology of the circuit conforms to the topology of FIG. 16.        The original design in the application note, is for a boost        converter. In the design of FIG. 17, the output voltage can be        lower than the peak voltage of the rectified AC mains or higher.        In contrast the boost converter of the application note can        produce only a voltage higher than the rectified AC mains peak        voltage.    -   The design of FIG. 17 does not need inrush current protection.        In contrast, a boost converter must charge the storage capacitor        to the peak of the rectified AC mains on power up and typically        needs inrush current protection.    -   The output of the power factor controller in FIG. 16 does not        share the same common terminal with the rectified AC mains or        the integrated circuit NCP1650. In FIG. 17F there is a reference        list of the four different common terminals used in the circuit        of FIG. 17.    -   The start-up circuit (FIG. 17B) has been modified because the        output voltage is not referred to the ground of the integrated        circuit.    -   The feedback circuit FIG. 17C) has been modified because the        output voltage is not referred to the ground of the integrated        circuit.    -   The circuit in FIG. 17F is optional. If the circuit is added, a        microcontroller can produce pseudo-random variations in the        frequency of the oscillator of the NCP1650 integrated circuit.        This is another example of the idea disclosed in FIG. 9I.    -   A generic DC-DC converter is added in FIG. 17B. This is an        example of the block (1607) in FIG. 16.

In FIG. 17A, a diode bridge (1701) rectifies the AC mains into a fallwave rectified AC waveform. The value of capacitor C1 (1702) has beenchosen so the voltage in the point V₁ of the circuit (the positiveoutput of the diode bridge rectifier (1701)) tracks the voltage producedby the rectifier. The circuit also includes capacitor (1711) that hasbeen placed in another side of the picture to show the fact thatcapacitor (1602) in FIG. 16, can be made of a number of capacitors inparallel. The position of the capacitors has to be considered withattention to EMI compatibility of the final product. All the capacitorin parallel with C1 (1702) (or (1602) in FIG. 16) must have good highfrequency characteristics. It must be noted that the output of thecircuit of FIG. 17A is the port from V₁ to V_(OP), consequently thepoint V₁ has been made the common for the DC-DC converter circuit inFIG. 17E.

Capacitor C3 (1707) is the storage capacitor of the power factorcontroller. It is in parallel with the output. The MOSFET MF1, (1709) inFIG. 17A is the switch (1605) of FIG. 16. When the MOSFET is conducting,the inductance L1 (1703) is connected across the input voltage andcurrent flows into it, growing linearly during the period of time theMOSFET is conducting (T_(ON)). At the frequency of commutation of thecircuit, the full wave rectified AC mains remains practically constantfor the period of commutation. The IC controller in FIG. 17, theNCP1650, of FIG. 17D is not referenced to the negative terminal of thediode bridge. Instead it is referred to the mid point between the MOSFET(1709) and the current sensing resistor R2 (1708) in FIG. 17A. The chipuses the negative voltage with respect to the common node of the chip'scircuit (pin 15 in FIG. 17D), as current information to decide when toturn of the MOSFET MP1 (1709).

When the MOSFET (1709) is off, diode D2 (1706) takes the currentcirculating from the inductance (1703). The current through theinductance can not change instantaneously and starts flowing into boththe load and the storage capacitor C3 (1707) connected in parallel withthe load. With the MOSFET MP1 non-conducting and the diode D2conducting. The inductance is just in parallel with the seriescombination the diode D2, itself in series with the parallel combinationof the capacitor C3 (1707) and the load (converter of FIG. 17B). If theMOSFET is kept non-conducting, current keeps flowing, charging capacitorC3 (the portion not taken by the load) until it reaches zero and thediode D2 becomes reverse biased from the voltage across the storagecapacitor C3.

If the MOSFET conducts before the current reaches zero, then the circuitis said to operate in the continuous current mode. The process startsagain in the next cycle. By the way the integrated circuit NCP1650(1733) controls the maximum value of the current trough the inductance,and the conduction time of the MOSFET, the filtered waveform of thetrain of triangular current pulses (or trapezoidal, in continuouscurrent mode) taken from capacitor C1 (1702) is filtered by the actionof the capacitor into a normally good approximation of full waverectified AC mains current waveform synchronous with the AC mains, thatyields close to unity power factor when “seen” from the AC mains side.

The inductance L1 has an additional winding that provides power for thechip itself. The inductance L1 works as the primary of a transformer.The diode D1 (1704) and the capacitor C2 (1705), provide rectificationand filtering of the wave in the secondary side of the arrangement.

FIG. 17B is the start up sub-circuit. The circuit is similar to theapplication note referenced above. The changes reflect the fact that theoutput can not bias the MOSFET to turn it off. Hence capacitors (1719)and (1720) has been added. Also resistor (1715) discharge capacitors(1719) and (1720) when the system is not powered. MOSFET MF2 (1717) ismade to conduct because of the voltage difference between V_(BIAS) (thatis low, at the start-up) and the voltage over the zener diode Z1 (1718).As V_(BIAS) approaches the working voltage the difference with the zenervalue is smaller and the MOSFET (1717) stops conducting. As the input ofthe start-up circuit is the full wave rectified AC mains (and not theoutput storage capacitor as in the original application note) withoutthe capacitors (1719) and (1720) the voltage over the zener diode ispulsating, (when the Input voltage of the full wave rectified AC mainsis lower than the zener value) and the MOSFET is periodically turned-on.Hence without the capacitors (1719) and (1720), the start-up circuitoverheat.

FIG. 17C is the feedback path. Those skilled in the art of switched modepower supplies will recognise the standard optically coupled feedbacknetwork formed with IC2 ((1730), typically a TL31 programmable zenerdiode IC. Resistors network (1731) formed by R15, RV1 and R14, set thevalue of the output voltage. When the output voltage grows, the voltagedrop across resistor R13 (1732) also grows. If this happens, the currenttrough the diode of the opto-coupler IC3 (1729) grows too. The outputtransistor of the opto-coupler conducts more and the voltage overresistor R11 (1728) grows. This voltage is directly feed to the input ofthe error comparator of the NCP1650 control loop (pin 6 of (1733)). Thenetwork form by resistor R8 (1725), R9 (1726) and diode (1727) has beenadded because the feedback network is not taken from a ground referencedstorage capacitor like in the original application not boost converter.The network formed by resistor R8 (1725), R9 (1726) and diode (1727)guarantee that there is a minimum voltage in the input of the errorcomparator of the NCP1650 control loop pin 6 of (1733)) when thereference voltage of the device is enabled after the power-up sequencefinishes. The network (1724) is for compensating the feedback loop. Thevalues depend on the type of load and the specifications of theconverter.

FIG. 17D is the controller IC itself, the NCP1650 (1733). All thecomponents in the circuit are mandatory as per the manufacturerspecifications. Refer to the data sheet of the device (ref. 26) and itsapplication notes (page 67 of reference 25).

FIG. 17D shows an example, in block diagram of the block (1607) in FIG.16. A classical push-pull converter has been chosen as an example only.Any type of DC-DC or DC to AC converter could be used in block (1607).Detailed operation and design methodologies of controller (17E02),MOSFET's (17E03) and (17E04), all components coupled with transformer(17E05), the output filtering network (7E06 to 17E10) and thetransformer (17E05) itself, can be found in the following references:

-   -   Reference 3, page 220.    -   Reference 7, page 116 and 153.    -   Reference 9, page 37.    -   Reference 10, pages 2.147 to 2.151 and 2.153 to 2.159.    -   Reference 11, pages 34 to 38.

The feedback network for the converter in FIG. 17E is based on the samecomponents and operating principle than the feedback network used inFIG. 17C. However, components R04, C03 and C02, (network 17E16) has beenadded foe completeness. The network (17E16) is used for dynamiccompensation of the feedback loop. In most of the circuits of this type,compensation is added around IC5 (17E13). (Unlike however the NCP1650that is designed so the compensation is added to a pin 7 of the chip(1724) in FIG. 17C). The graphic in FIG. 17E shows the full waverectified AC mains in the common point of the circuit in the primaryside of transformer (17E05). The reference point of the secondary sidecan be chosen independently of any other point in the circuit (providethat the isolation given by the specifications of the transformer isadequate). In FIG. 17E, earth has been chosen for the common point (or“ground terminal”) of the output of converter (1607).

FIG. 17F shows a list of the four different common terminals symbolsused in the circuit. FIG. 17F also shows an example of how theoscillator of NCP1650 can be frequency modulated by an external circuit.A microcontroller producing a pulse width modulated waveform in thediode terminals of the opto-coupler IC06 (17F08) can producepseudo-random variations in the frequency of the oscillator of theNCP1650 integrated circuit. This is another example of the ideadisclosed in FIG. 9I.

When current flows though the LED of the opto-coupler IC06 (17F08) theoutput transistor conducts and the pulse width modulated waveform isreproduced over the resistor R105 (17F10). Resistor R104 (17F09) is usedto drop the voltage of the VREF supply by forming a resistive dividerwith resistor R105 (17F10). Resistor R103 (17F07) and capacitor C100(17F06) filter the pulse width modulated signal and its low frequencycontent is available in the non inverting input of the operationalamplifier ICS (17F04). The operational amplifier IC5 (17F04) itsfeedback network (R101 (17F03), R100 (17F01)) and the P channel MOSFETMF5 (17F02), form a regulated current source in parallel with thecurrent source of 200 micro-amperes provided by pin 14 of the CP1650 IC,to charge the timing capacitor CT (1738). Resistor R102 (17F05) isincluded for stability. If R101 is made equal to several hundred timesthe value of R100, the current out of the drain terminal of the Pchannel MOSFET MF5 (17F02) will be the voltage difference betweenV^(REF) and the voltage at the non-inverting input of the op-amp IC5(17F04) divided by the value of resistance R100 (17F01). In this way,the additional current charging the timing capacitor CT (1738), willvaried with the low frequency content of the pulse width modulatedsignal at the input of the opto-coupler.

While only a limited number of embodiments have been disclosed, numerousmodifications and substitutions can be made without departing from thescope and spirit of the invention.

REFERENCES

-   1.—Brush-loss Permanent Magnet Motor Design    -   Duane C. Hanselman    -   McGraw Hill Inc 1994    -   ISBN 0-07-026025-7-   2.—A250 Watt Current—Controlled SMPS with Synchronous Rectification    -   By R. Pearce and D. Grant    -   Application note 960A. International Rectifier    -   Page 137 of    -   HEXFET Designer's Manual Volume I,    -   HDM-1 first printing, International rectifier 1993-   3.—Power Electronics    -   Converters, Applications, and Design    -   N. Mohan, T. M. Undeland and W. P. Robbins    -   John Wiley & Sons 1989    -   ISBN 0-471-61342-8-   4.—Motorola Application Note AN-876    -   Using Power MOSFET's in Stepping Motor Control    -   Published in the proceedings of Powercon 9, 1982-   5.—UC1637/2637/3637 Switched Mode Controller for DC Motor Drive data    sheet    -   Unitrode Integrated Circuits Corp. (Texas Instruments        Incorporated)-   6.—UC1637/2637/3637 Switched Mode Controller for DC Motor Drive    -   Application note U-102    -   Unitrode Integrated Circuits Corp. (Texas Instruments        Incorporated)-   7—Design of Solid States Power Supplies    -   Third Edition    -   Hnatek, E. P,    -   Van Nostrand Reinhold Co. 1989    -   ISBN 0-442-20768-9-   8.—. EMC for Product Designers    -   Meeting the European EMC Directive    -   Tim Williams    -   Butterworth Heinemann 1992    -   ISBN 0-7506-1264-9-   9.—Switching Power Supply Design    -   Abraham I. Pressman    -   McGraw-Hill, Inc 1991    -   ISBN 0-07-050806-2-   10.—Switch Mode Power Supply Handbook    -   Keith Billings    -   McGraw-Hill, Inc 1989    -   ISBN 0-07-005330-8-   11.—Practical Switching Power Supply Design    -   Marty Brown    -   Academic Press Inc 1990    -   ISBN 0-12-137030-5-   12.—Power Supply Cookbook (second edition)    -   Marty Brown    -   Butterworth Heinemann 2001    -   ISBN 0-7506-7329-X-   13.—An Economic Motor Drive With Very Few Components    -   SGS-Thomson Microelectronics Application Note    -   AN282/0589    -   Page 305, Designer's Guide to Power Products Application Manual        2^(nd) Edition, June 1992    -   SGS-Thomson Microelectronics-   14.—Designing With The L296 Monolithic Power Switching Regulator    -   Page 29/42, FIG. 36 (use as a motor speed controller)    -   SGS-Thomson Microelectronics Application Note    -   AN244/1288    -   Page 463 of Designer's Guide to Power Products Application        Manual    -   SGS-Thomson Microelectronics 2^(nd) Edition, June 1992-   15.—How to Drive DC Motors With Smart Power IC's    -   By Herbert Sax    -   SGS-Thomson Microelectronics Application Note    -   AN 380/0591    -   Page 215 of Designer's Guide to Power Products Application        Manual    -   SGS-Thomson Microelectronics 2^(nd) Edition, June 1992-   16.—Load Current Sensing in Switch-mode Bridge Motor Driving    Circuits    -   By Herbert Sax    -   SGS-Thomson Microelectronics Application Note    -   AN 452/0392    -   Page 231 of Designer's Guide to Power Products Application        Manual    -   SGS-Thomson Microelectronics 2^(nd) Edition, June 1992-   17.—Driving DC Motors    -   By Maiocchi    -   SGS-Thomson Microelectronics Application Note    -   AN 281/0189    -   Page 255 of Designer's Guide to Power Products Application        Manual    -   SGS-Thomson Microelectronics 2^(nd) Edition, June 1992-   18.—Switched-mode Drives for DC Motors    -   By Lester J. Hadley, Jr.    -   Philips Semiconductors Corporation, Application note    -   AN1221, December 1988-   19.—L292 Switch-mode Driver for DC motors    -   L292 Integrated circuit data sheet    -   SGS-Thomson Microelectronics, March 1993-   20.—LT1248 Power Factor Controller    -   LT1248 data sheet    -   Page 4/194 of 1994 Linear Data Book Volume III    -   Linear Technology Corporation-   21.—Unity Power Factor Power Supply    -   By B. Wilkinson and J. Mandelcom    -   U.S. Pat. No. 4,677,366    -   Jun. 30, 1987-   22.—MC33035 Brushless DC Motor Controller (MC33035 data sheet)    -   Publication Order Number: MC33035/D, April, 2001—Rev. 4-   Page 3050 in:    -   Analog Integrated Circuits, DL128/D    -   Power Management, Signal Conditioning and ASSP Devices    -   Rev. 7, March-2002 On Semiconductor-   23.—UC2845A Current Mode PWM controller    -   Data sheet    -   Unitrode Integrated Circuits Corp. (Texas Instruments        Incorporated)-   24.—Power Semiconductor Applications    -   Philips Components    -   Multiple authors,    -   SCD2, date of release April 1991 document number 9398 651 40011-   25.—Power Factor Correction Handbook    -   On Semiconductor    -   Multiple authors,    -   Document Number HBD853/D Rev. 1, June 2004-   26.—Data Sheet NCP1650 Power Factor Controller    -   On Semiconductor    -   Document Number NCP1650/D Rev. 8, August 2003-   27.—Data Sheet NCP1651 Single Stage Power Factor Controller    -   On Semiconductor    -   Document Number NCP1651/D Rev. 5, October 2003

1-46. (canceled)
 47. A system for driving a direct-current (DC) motorunder conditions of controlled DC current, from a DC voltage source of avalue larger than said motor operating voltage, independently of theoperating voltage of said motor, said system comprising: a pair of nodesfor connection of said DC motor, said nodes to be referred herein as thefirst node and the second node; said second node connected to a commonelectrical terminal of the system through an electrical path with lowimpedance, including low impedance to DC current; said DC motor isconnected between said first node and said second node; an inductiveelement to store energy and to act as a current source for said DCmotor, said inductive element is external to said DC motor, and not partof said DC motor main magnetic circuit, said inductive element isconnected to said first node, in series with said DC motor, saidinductive element being capable of operating in a buck converter at thepower level required to operate said DC motor and at the frequency ofcommutation of a first switch, the terminal of said inductive elementnot connected to said first node to be connected to a third node; saidfirst switch is connected to said inductive element at the third node,the terminal of said inductive element remote from said DC motor; saidfirst switch used for connected and disconnecting said inductive elementto a direct current (DC) voltage source; a terminal of said DC voltagesource not connected to said first switch, to be connected to saidcommon electrical terminal of the system; said first switch is acontrolled switch capable of being turned off and on (switch open orclosed) by control signals from a control system; said control systemoperates based on an error signal and a value of the desired operatingcurrent for said DC motor set externally to the system; the object ofsaid control system is to turn said first switch off and on (switch openor closed) in order to minimise said error signal and to keep theoperating current of said DC motor at said desired value; said firstswitch is a single pole switch; a second switch connected between saidthird node and said common electrical terminal of the system, parallelwith a combination of said inductive element and said DC motor arrangedin series; said second switch controlled so that a current circulatingthrough said inductive element circulates through said second switch ifsaid first switch is turned off (switch open) and disconnects saidinductive element from said DC voltage source; said second switch is asingle pole switch; a capacitor arranged for connection in parallel withsaid DC motor to limit a resulting voltage over said DC motor, saidcapacitor being capable of operating in a buck converter at a powerlevel required to operate said DC motor and at the frequency ofcommutation of said first switch, said capacitor is connected betweensaid first node and a low impedance path to said common electricalterminal of the circuit; a current sensor for measuring a currentthrough said DC motor; the output of said current sensor connected tosaid control system of said first switch to generate said error signalfor the operation of said control system controlling the operation ofsaid first switch, means for controlling operation of said second switchdependent upon the state of the first switch.
 48. The system accordingto claim 47, wherein the voltage of said DC voltage source is largerthan the nominal rated voltage of said DC motor.
 49. A system fordriving a direct-current (DC) motor under conditions of controlled DCcurrent, independently of the operating voltage of said motor, saidsystem comprising: a pair of nodes for connection of said DC motor, saidnodes to be referred to herein as the first node and the second node;said second node connected to a common electrical terminal of the systemthrough an electrical path with low impedance, including low impedanceof DC current; said DC motor is connected between said first node andsaid second node; a capacitor arranged for connection in parallel withsaid motor to limit a resulting voltage over said motor, one terminal ofsaid capacitor connected to said first node the other terminal of saidcapacitor to be connected through a low impedance to said commonterminal of the system, said capacitor being capable of operating in abuck converter at the power level required to operate said DC motor andat the frequency of commutation of a first switch; an inductive elementwith one terminal connected to said common terminal of the systemthrough a low impedance path, the other terminal of said inductiveelement, referred to herein as the third node, is connected to saidfirst switch; said inductive element is used to store energy and to actas a current source for said DC motor, said inductive element beingexternal to said DC motor, and not part of the said DC motor mainmagnetic circuit, said inductive element being capable of operating in abuck converter at the power level required to operate said DC motor andat the frequency of commutation of said first switch; said first switchis connected to said inductive element in the third node, said firstswitch used for connecting and disconnecting said third node to a DCvoltage source; a terminal of said DC voltage source not connected tosaid first switch, to be connected to said common electrical terminal ofthe system; said first switch being a controlled switch capable of beingturned off and on (switch open and closed) by control signals from acontrol system; said control system operates based on an error signaland a value of a desired operating current for said DC motor, setexternally to the system; the object of said control system is to turnsaid first switch off and on (switch open and closed) in order tominimise said error signal and to keep the operating current of said DCmotor at said desired value; said first switch is a single pole switch;a second switch connected between said first node and said third node,that is, in series with the parallel combination of said motor and saidcapacitor, and connected to the common node between the first switch andsaid inductive element; said second switch controlled so that a currentcirculating through said inductive element circulates through saidsecond switch if the first switch is turned off (switch opened) anddisconnects the third node from said DC voltage source; said secondswitch is a single pole switch; a current sensor for measuring a currentthrough said DC motor; the output of said current sensor connected tosaid control system of said first switch to generate said error signalfor the operation of said control system, controlling the operation ofsaid first switch, and means for controlling operation of said secondswitch dependent upon the state of the first switch.
 50. A system fordriving a direct-current (DC) motor under conditions of controlled DCcurrent, independently of the operating voltage of said motor, saidsystem comprising: a pair of nodes for connection of said DC motor, saidnodes to be referred to herein as the first node and the second node,said DC motor is connected between said first and said second node; acapacitor arranged for connection in parallel with said motor, betweensaid first node and said second node, to limit a resulting voltage oversaid motor, said first node, connected to a terminal of said capacitorand said motor, being also connected to a DC voltage source, saidcapacitor being capable of operating in a buck converter at the powerlevel required to operate said DC motor and at the frequency ofcommutation of a first switch; the other terminal of said DC voltagesource to be connected to a common electrical terminal of the system; aninductive element with one terminal connected to said first node, acommon node of said DC voltage source, said capacitor and said DC motor,the other terminal of said inductive element, referred to herein as thethird node and is connected to said first switch; said inductive elementused to store energy and to act as a current source for said DC motor,said inductive element being external to said DC motor, and not part ofthe said DC motor main magnetic circuit, said inductive element beingcapable of operating in a buck converter at the power level required tooperate said DC motor and at the frequency of commutation of said firstswitch; said first switch is connected to said inductive element in thethird node, the other terminal of said first switch, not connected tothe third node is connected to said common electrical terminal of thesystem through an electrical path with low impedance, including lowimpedance to DC current; said first switch used for connecting anddisconnecting the third node to said common electrical terminal of thesystem; said first switch being a controlled switch capable of beingturned off and on (switch open and closed) by control signals from acontrol system; said control system operates based on an error signaland a value of a desired operating current for said DC motor setexternally to the system; the object of said control system is to turnsaid first switch off and on (switch open and closed) in order tominimise said error signal and to keep the operating current of said DCmotor at said desired value; said first switch is a single pole switch;a second switch connected between said second and said third node, saidsecond switch controlled so that a current circulating through saidinductive element circulates through said second switch if the firstswitch is turned off (switch opens) and disconnects the third node fromsaid common electrical terminal of the system; said second switch is asingle pole switch; a current sensor for measuring a current throughsaid DC motor; the output of said current sensor connected to saidcontrol system of said first switch to generate said error signal forthe operation of said control system, controlling the operation of saidfirst switch; and means for controlling operation of said second switchdependent upon the state of the first switch.
 51. The system accordingto claim 47, wherein said second switch is a diode connected withappropriate polarity so that current circulating through said inductiveelement circulates through said diode if said first switch is open,disconnecting said inductive element.
 52. The system according to claim47, wherein said second switch is an electronic switch for synchronousrectification connected with appropriate polarity so that currentcirculating through said inductive element circulates through saidelectronic switch if said first switch is open, disconnecting saidinductive element.
 53. The system according to claim 47, wherein saidfirst switch is an electronic switch.
 54. A system for driving adirect-current (DC) motor under conditions of a controlled averagecurrent, a voltage of a DC power supply having a larger or smaller valuethan a motor nominal voltage, said system comprising: an inductiveelement for connection in series with the DC motor, said inductiveelement capable of operating in a buck converter at the power levelrequired to operate said DC motor and at the frequency of a firstswitch; an arrangement including a plurality of switches, diodes and amagnetic system, said arrangement coupled to said inductive element forconnecting and disconnecting a terminal of said inductive element remotefrom said motor to a voltage source, said arrangement configured ascircuit selected from the group consisting of: a forward DC-DC convertera push-pull DC-DC converter a half-bridge DC-DC converter adiagonal-half bridge DC-DC converter a full bridge DC-DC converter acapacitor arranged for connection in parallel with said motor to limit aresulting voltage over the motor, said capacitor capable of operating ina buck converter at the power level required to operate said DC motorand at the frequency of commutation of said first switch; means formeasuring a current through the motor; and means for controllingoperation of said arrangement dependent upon said measured current inthe motor.
 55. The system according to claim 47, wherein said inductiveelement is an inductor, or a winding of a transformer.
 56. The systemaccording to claim 47, wherein a current through the inductive elementcan be controlled independently from a current through the motor, thebalance of electrical charge being accumulated or taken from thecapacitor in parallel with the motor.
 57. The system according to claim47, wherein a current through the inductive element is modulated as afull wave rectified sinusoid synchronous with the AC main voltage sothat the power factor of the system, as a load to the AC main isimproved.
 58. A system for driving a direct-current (DC) motor underconditions of a controlled average current, a voltage of a DC powersupply having a larger or smaller value than a motor nominal voltage,said system comprising: a diode; a magnetic transformer connected inseries with said diode in a circuit arrangement selected from the groupconsisting of a flyback DC-DC converter and a ringing choke DC-DCconverter, said transformer and said diode for connection in series withthe DC motor; a switch coupled to said magnetic transformer and saiddiode for connecting and disconnecting a terminal of said magnetictransformer and said diode remote from said motor to a voltage source; acapacitor arranged for connection in parallel with said motor to limit aresulting voltage over the motor, said capacitor capable of operating ina buck converter at the power level required to operate said DC motorand at frequency of commutation of said switch; means for measuring acurrent through the motor; and means for controlling operation of saidswitch dependent upon said measured current in the motor.
 59. The systemaccording to claim 58, wherein said switch is an electronic switch. 60.A system for driving a direct-current (DC) motor under conditions of acontrolled average current, a voltage of a DC power supply having alarger or smaller value than a motor nominal voltage, said systemcomprising: an electronic synchronous rectification switch; a magnetictransformer connected in series with said synchronous rectificationswitch in a circuit arrangement selected from the group consisting of aflyback DC-DC converter and a ringing choke DC-DC converter, saidtransformer and said synchronous rectification switch for connection inseries with the DC motor; a switch coupled to said magnetic transformerand said synchronous rectification switch for connecting anddisconnecting a terminal of said magnetic transformer and saidsynchronous rectification switch remote from said motor to a voltagesource; a capacitor arranged for connection in parallel with said motorto limit a resulting voltage over the motor; means for measuring acurrent through the motor; and means for controlling operation of saidswitch dependent upon said measured current in the motor.
 61. The systemaccording to claim 60, wherein said switch is an electronic switch. 62.The system according to claim 58, wherein a current through the flybackinductance can be controlled independently from a current through themotor, the balance of electrical charge being accumulated or taken fromthe capacitor in parallel with the motor.
 63. The system according toclaim 58, wherein a current through the flyback inductance is modulatedas a full wave rectified sinusoid synchronous with the AC main voltageso that the power factor of the system, as a load to the AC main isimproved.
 64. The system according to claim 47, further comprising a DCmotor.
 65. The system according to claim 64, wherein said DC motorincludes a brush-less DC motor.
 66. The system according to claim 65,wherein said DC motor includes an electronic commutator for saidbrush-less DC motor.
 67. The system according to claim 47, wherein saidmeans for measuring said current through the motor includes means forcalculating said current through the motor dependent upon currentmeasured in another part of said system.
 68. The system according toclaim 47, wherein a frequency of a pulse width modulated waveform,resulting from operation of said switches, is randomized to facilitateEMI compliance.
 69. An airflow apparatus, comprising: a brush-less DCmotor; an electronic circuit for controlling operation of saidbrush-less DC motor; a power supply for said electronic circuit separatefrom a power supply for said brush-less DC motor, said power supply forsaid electronic circuit adapted to use a voltage resulting from saidbrush-less DC motor in operation once said resulting voltage reaches asuitable value; and means for reducing power to said electronic circuitfrom said power supply once said resulting voltages reaches saidsuitable value.
 70. The airflow apparatus according to claim 69, whereinsaid apparatus is an airflow generator.
 71. The airflow apparatusaccording to claim 70, wherein said airflow generator is for use inmedical applications.
 72. The airflow apparatus according to claim 69,wherein said reducing means comprises means for disconnecting saidelectronic circuit from said power supply for said control electronics.73. The airflow apparatus according to claim 69, wherein said electroniccircuit comprises an electronic commutator or driving electronics forsaid brush-less DC motor.
 74. The airflow apparatus according to claim69, wherein said electronic circuit comprises a buck converter or downconverter switched mode power supply connected to a rectified AC mainvoltage.
 75. The airflow apparatus according to claim 69, wherein saidelectronic circuit comprises means for controlling current through saidbrush-less DC motor.
 76. The airflow apparatus according to claim 69,wherein said electronic circuit utilizes a pulse width modulated squarewave applied through a transformer to control a voltage over said motor.77. The airflow apparatus according to claim 76, wherein saidtransformer is part of a forward converter, a push-pull converter, ahalf bridge converter, a diagonal half bridge converter, a bridgeconverter, or a flyback converter.
 78. The airflow apparatus accordingto claim 69, further comprising a driving system wherein said electroniccircuit controls a current through said brush-less DC motor (BLDCM),where said inductive element is a winding of a transformer having aplurality of secondary windings, a secondary winding being sued toprovide power to said electronic circuit.
 79. The airflow apparatusaccording to claim 69, wherein said airflow generator is for use as acooling fan or a ventilation fan.
 80. The airflow apparatus according toclaim 79, wherein a plurality of brush-less DC motor-driven ventilationfans or cooling fans are connected in series between each other.
 81. Asystem for intermittently powering a microprocessor based system from aDC voltage higher than the voltage required by the system to operate,comprising: a capacitor; means to charge said capacitor from the DCvoltage with a current substantially smaller than the current themicroprocessor based system needs to operate; a switch coupled to saidcapacitor so that said switch can connect power to the microprocessorbased system from the charge accumulated in the capacitor; means forsensing the voltage in the capacitor and causing the switch to closeonce the voltage in said capacitor reaches a desired value; and meansfor keeping the switch closed while the voltage in said capacitor isover a desired value, but less than the value that caused said sensingmeans to close the switch.
 82. The system according to claim 81, whereinsaid switch is an electronic switch.
 83. The system according to claim81, further comprising means for limiting current through said switch.84. The system according to claim 56 wherein the current through themotor is calculated from the variation of the voltage across thecapacitor in parallel with the motor.
 85. The system according to claim57, wherein the instant in which the sinusoidal waveform of the AC maincrosses zero is sensed to synchronise the modulation performed to thecurrent through the inductive element with the waveform in the AC main.86. The system according to claim 47, wherein the voltage over the DCmotor is used to estimate the speed of the motor.
 87. A switching basedalternating current (AC) to direct current (DC) converter, comprising: arectifier adapted to be connected to an alternating current (AC) mainsline, said rectifier having at least one output comprising two nodes, tobe referred to herein as the output common node and the input commonnode; a first capacitor for noise reduction connected between saidoutput common node and said input common node; an inductive elementconnected to said output common node; a first switch connected betweensaid input common node and the terminal of said inductive element notconnected to said output common node; said first switch used forconnecting and disconnecting a terminal of said inductive element remotefrom said output common node, the connected node between said inductiveelement and said first switch to be referred to herein as the firstnode; a second switch connected to said first node, controlled so thatthe current circulating through said inductive element circulatesthrough said second switch when said first switch disconnects saidinductive element from said input common node, the node of the secondswitch not connected to said first node to be referred to herein as theDC output node; a second capacitor for energy storage connected betweensaid DC output node and said output common node; means for sensing acurrent through said inductive element; means for sensing the voltageacross said first capacitor; means for sensing the voltage across saidsecond capacitor, the voltage across said second capacitor to bereferred to herein as the output voltage; a control circuit using thesensed value of the voltage across said first capacitor, and the sensedvalue of the output voltage, said control circuit connected to saidfirst switch to maintain said output voltage between defined limits byoperating said first switch in a way that said current through saidinductive element tracks the waveform of the alternating current linevoltage, to cause said switching based AC to DC converter to exhibitunity power factor to the alternating current line.
 88. The converterdefined in claim 87, wherein said second switch is a diode.
 89. Theconverter defined in claim 87, wherein said first switch comprises afield effect transistor.
 90. The converter defined in claim 87, whereinsaid rectifier comprises a full-wave diode rectifier.
 91. The converterdefined in claim 87, wherein said inductive element is an inductor, or awinding of a transformer.
 92. The converter defined in claim 87, whereina frequency of a pulse width modulated waveform controlling said firstswitch or said second switch, resulting from operation of said switches,is randomized to facilitate EMI compliance.